Lecture Notes on Mobile Communication

upload by Avadhootha Rams


Lecture Notes on Mobile Communication
A Curriculum Development Cell Project Under QIP, IIT Guwahati
Dr. Abhijit Mitra
Department of Electronics and Communication Engineering
Indian Institute of Technology Guwahati
Guwahati { 781039, India
November 2009
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Preface
It's been many years that I'm teaching the subject \Mobile Communication"
(EC632) to the IIT Guwahati students and the current lecture notes intend to act
as a supplement to that course so that our students can have an access to this
course anytime. This course is mainly aimed toward senior year students of the
ECE discipline, and in particular, for the nal year BTech, rst year MTech and
PhD students. However, this does not necessarily imply that any other discipline
students can not study this course. Rather, they also should delve deeper into
this course since mobile communication is a familiar term to everyone nowadays.
Although the communication aspects of this subject depends on the fundamentals of
another interesting subject, communication engineering, I would strongly advocate
the engineering students to go through the same in order to grow up adequate
interest in this eld. In fact, the present lecture notes are designed in such a way
that even a non-ECE student also would get certain basic notions of this subject.
The entire lecture notes are broadly divided into 8 chapters, which, I consider to
be most rudimentary topics to know the basics of this subject. The advance level
topics are avoided intensionally so as to give space to the possibility of developing
another lecture note in that area. In fact, this area is so vast and changing so fast
over time, there is no limit of discussing the advanced level topics. The current focus
has been therefore to deal with those main topics which would give a senior student
su cient exposure to this eld to carry out further study and/or research. Initially,
after dealing with the introductory concepts (i.e., what is mobile communication,
how a mobile call is made etc) and the evolution of mobile communication systems till
the present day status, the cellular engineering fundamentals are discussed at length
to make the students realize the importance of the practical engineering aspects of
this subject. Next, the di erent kinds of mobile communication channels is taken
up and large scale path loss model as well as small scale fading e ects are dealt,
both with simulation and statistical approaches. To enhance the link performance
amidst the adverse channel conditions, the transmitter and receiver techniques are
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discussed next. It is further extended with three main signal processing techniques
at the receiver, namely, equalization, diversity and channel coding. Finally, di erent
kinds of multiple access techniques are covered at length with the emphasis on how
several mobile communication techniques evolve via this. It should also be mentioned
that many gures in the lecture notes have been adopted from some standard text
books to keep the easy
ow of the understanding of the topics.
During the process of developing the lecture notes, I have received kind helps
from my friends, colleagues as well as my post graduate and doctoral students which
I must acknowledge at the onset. I'm fortunate to have a group of energetic students
who have helped me a lot. It is for them only I could nish this project, albeit a
bit late. My sincere acknowledgment should also go to my parents and my younger
brother who have nicely reciprocated my oblivion nature by their nourishing and
generous attitude toward me since my childhood.
Finally, about the satisfaction of the author. In general, an author becomes
happy if he/she sees that his/her creation could instill certain sparks in the reader's
mind. The same is true for me too. Once Bertrand Russell said \Science may set
limits to knowledge, but should not set limits to imagination". If the readers can
visualize the continuously changing technology in this eld after reading this lecture
notes and also can dream about a future career in the same, I'll consider my endeavor
to be successful. My best wishes to all the readers.
Abhijit Mitra
November 2009
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Contents
1 Introductory Concepts 1
1.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
1.2 Evolution of Mobile Radio Communications . . . . . . . . . . . . . . 1
1.3 Present Day Mobile Communication . . . . . . . . . . . . . . . . . . 3
1.4 Fundamental Techniques . . . . . . . . . . . . . . . . . . . . . . . . . 4
1.4.1 Radio Transmission Techniques . . . . . . . . . . . . . . . . . 5
1.5 How a Mobile Call is Actually Made? . . . . . . . . . . . . . . . . . 7
1.5.1 Cellular Concept . . . . . . . . . . . . . . . . . . . . . . . . . 7
1.5.2 Operational Channels . . . . . . . . . . . . . . . . . . . . . . 8
1.5.3 Making a Call . . . . . . . . . . . . . . . . . . . . . . . . . . 8
1.6 Future Trends . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
1.7 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
2 Modern Wireless Communication Systems 11
2.1 1G: First Generation Networks . . . . . . . . . . . . . . . . . . . . . 11
2.2 2G: Second Generation Networks . . . . . . . . . . . . . . . . . . . . 11
2.2.1 TDMA/FDD Standards . . . . . . . . . . . . . . . . . . . . . 12
2.2.2 CDMA/FDD Standard . . . . . . . . . . . . . . . . . . . . . 12
2.2.3 2.5G Mobile Networks . . . . . . . . . . . . . . . . . . . . . . 12
2.3 3G: Third Generation Networks . . . . . . . . . . . . . . . . . . . . . 13
2.3.1 3G Standards and Access Technologies . . . . . . . . . . . . . 14
2.3.2 3G W-CDMA (UMTS) . . . . . . . . . . . . . . . . . . . . . 14
2.3.3 3G CDMA2000 . . . . . . . . . . . . . . . . . . . . . . . . . . 16
2.3.4 3G TD-SCDMA . . . . . . . . . . . . . . . . . . . . . . . . . 18
2.4 Wireless Transmission Protocols . . . . . . . . . . . . . . . . . . . . 19
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2.4.1 Wireless Local Loop (WLL) and LMDS . . . . . . . . . . . . 19
2.4.2 Bluetooth . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
2.4.3 Wireless Local Area Networks (W-LAN) . . . . . . . . . . . . 20
2.4.4 WiMax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
2.4.5 Zigbee . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
2.4.6 Wibree . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
2.5 Conclusion: Beyond 3G Networks . . . . . . . . . . . . . . . . . . . . 22
2.6 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
3 The Cellular Engineering Fundamentals 23
3.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
3.2 What is a Cell? . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
3.3 Frequency Reuse . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
3.4 Channel Assignment Strategies . . . . . . . . . . . . . . . . . . . . . 27
3.4.1 Fixed Channel Assignment (FCA) . . . . . . . . . . . . . . . 27
3.4.2 Dynamic Channel Assignment (DCA) . . . . . . . . . . . . . 27
3.5 Hando Process . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
3.5.1 Factors In
uencing Hando s . . . . . . . . . . . . . . . . . . 29
3.5.2 Hando s In Di erent Generations . . . . . . . . . . . . . . . 31
3.5.3 Hando Priority . . . . . . . . . . . . . . . . . . . . . . . . . 33
3.5.4 A Few Practical Problems in Hando Scenario . . . . . . . . 33
3.6 Interference & System Capacity . . . . . . . . . . . . . . . . . . . . . 34
3.6.1 Co-channel interference (CCI) . . . . . . . . . . . . . . . . . . 34
3.6.2 Adjacent Channel Interference (ACI) . . . . . . . . . . . . . . 37
3.7 Enhancing Capacity And Cell Coverage . . . . . . . . . . . . . . . . 38
3.7.1 The Key Trade-o . . . . . . . . . . . . . . . . . . . . . . . . 38
3.7.2 Cell-Splitting . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
3.7.3 Sectoring . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43
3.7.4 Microcell Zone Concept . . . . . . . . . . . . . . . . . . . . . 46
3.8 Trunked Radio System . . . . . . . . . . . . . . . . . . . . . . . . . . 47
3.9 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53
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4 Free Space Radio Wave Propagation 54
4.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54
4.2 Free Space Propagation Model . . . . . . . . . . . . . . . . . . . . . 55
4.3 Basic Methods of Propagation . . . . . . . . . . . . . . . . . . . . . . 57
4.3.1 Re
ection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 57
4.3.2 Di raction . . . . . . . . . . . . . . . . . . . . . . . . . . . . 58
4.3.3 Scattering . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 58
4.4 Two Ray Re
ection Model . . . . . . . . . . . . . . . . . . . . . . . . 59
4.5 Di raction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 63
4.5.1 Knife-Edge Di raction Geometry . . . . . . . . . . . . . . . . 64
4.5.2 Fresnel Zones: the Concept of Di raction Loss . . . . . . . . 66
4.5.3 Knife-edge di raction model . . . . . . . . . . . . . . . . . . . 68
4.6 Link Budget Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . 69
4.6.1 Log-distance Path Loss Model . . . . . . . . . . . . . . . . . 69
4.6.2 Log Normal Shadowing . . . . . . . . . . . . . . . . . . . . . 70
4.7 Outdoor Propagation Models . . . . . . . . . . . . . . . . . . . . . . 70
4.7.1 Okumura Model . . . . . . . . . . . . . . . . . . . . . . . . . 70
4.7.2 Hata Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . 71
4.8 Indoor Propagation Models . . . . . . . . . . . . . . . . . . . . . . . 72
4.8.1 Partition Losses Inside a Floor (Intra-
oor) . . . . . . . . . . 72
4.8.2 Partition Losses Between Floors (Inter-
oor) . . . . . . . . . 73
4.8.3 Log-distance Path Loss Model . . . . . . . . . . . . . . . . . 73
4.9 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 73
4.10 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 73
5 Multipath Wave Propagation and Fading 75
5.1 Multipath Propagation . . . . . . . . . . . . . . . . . . . . . . . . . . 75
5.2 Multipath & Small-Scale Fading . . . . . . . . . . . . . . . . . . . . 75
5.2.1 Fading . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 76
5.2.2 Multipath Fading E ects . . . . . . . . . . . . . . . . . . . . 76
5.2.3 Factors In
uencing Fading . . . . . . . . . . . . . . . . . . . . 76
5.3 Types of Small-Scale Fading . . . . . . . . . . . . . . . . . . . . . . . 77
5.3.1 Fading E ects due to Multipath Time Delay Spread . . . . . 77
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5.3.2 Fading E ects due to Doppler Spread . . . . . . . . . . . . . 78
5.3.3 Doppler Shift . . . . . . . . . . . . . . . . . . . . . . . . . . . 79
5.3.4 Impulse Response Model of a Multipath Channel . . . . . . . 80
5.3.5 Relation Between Bandwidth and Received Power . . . . . . 82
5.3.6 Linear Time Varying Channels (LTV) . . . . . . . . . . . . . 84
5.3.7 Small-Scale Multipath Measurements . . . . . . . . . . . . . . 85
5.4 Multipath Channel Parameters . . . . . . . . . . . . . . . . . . . . . 87
5.4.1 Time Dispersion Parameters . . . . . . . . . . . . . . . . . . 87
5.4.2 Frequency Dispersion Parameters . . . . . . . . . . . . . . . . 89
5.5 Statistical models for multipath propagation . . . . . . . . . . . . . . 90
5.5.1 NLoS Propagation: Rayleigh Fading Model . . . . . . . . . . 91
5.5.2 LoS Propagation: Rician Fading Model . . . . . . . . . . . . 93
5.5.3 Generalized Model: Nakagami Distribution . . . . . . . . . . 94
5.5.4 Second Order Statistics . . . . . . . . . . . . . . . . . . . . . 95
5.6 Simulation of Rayleigh Fading Models . . . . . . . . . . . . . . . . . 96
5.6.1 Clarke's Model: without Doppler E ect . . . . . . . . . . . . 96
5.6.2 Clarke and Gans' Model: with Doppler E ect . . . . . . . . . 96
5.6.3 Rayleigh Simulator with Wide Range of Channel Conditions 97
5.6.4 Two-Ray Rayleigh Faded Model . . . . . . . . . . . . . . . . 97
5.6.5 Saleh and Valenzuela Indoor Statistical Model . . . . . . . . 98
5.6.6 SIRCIM/SMRCIM Indoor/Outdoor Statistical Models . . . . 98
5.7 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 99
5.8 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 99
6 Transmitter and Receiver Techniques 101
6.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 101
6.2 Modulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 101
6.2.1 Choice of Modulation Scheme . . . . . . . . . . . . . . . . . . 102
6.2.2 Advantages of Modulation . . . . . . . . . . . . . . . . . . . . 102
6.2.3 Linear and Non-linear Modulation Techniques . . . . . . . . . 103
6.2.4 Amplitude and Angle Modulation . . . . . . . . . . . . . . . 104
6.2.5 Analog and Digital Modulation Techniques . . . . . . . . . . 104
6.3 Signal Space Representation of Digitally Modulated Signals . . . . . 104
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6.4 Complex Representation of Linear Modulated Signals and Band Pass
Systems . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 105
6.5 Linear Modulation Techniques . . . . . . . . . . . . . . . . . . . . . 106
6.5.1 Amplitude Modulation (DSBSC) . . . . . . . . . . . . . . . . 106
6.5.2 BPSK . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 107
6.5.3 QPSK . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 107
6.5.4 O set-QPSK . . . . . . . . . . . . . . . . . . . . . . . . . . . 108
6.5.5 =4 DQPSK . . . . . . . . . . . . . . . . . . . . . . . . . . . 110
6.6 Line Coding . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 110
6.7 Pulse Shaping . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 111
6.7.1 Nyquist pulse shaping . . . . . . . . . . . . . . . . . . . . . . 112
6.7.2 Raised Cosine Roll-O Filtering . . . . . . . . . . . . . . . . 113
6.7.3 Realization of Pulse Shaping Filters . . . . . . . . . . . . . . 113
6.8 Nonlinear Modulation Techniques . . . . . . . . . . . . . . . . . . . . 114
6.8.1 Angle Modulation (FM and PM) . . . . . . . . . . . . . . . . 114
6.8.2 BFSK . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 116
6.9 GMSK Scheme . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 118
6.10 GMSK Generator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 119
6.11 Two Practical Issues of Concern . . . . . . . . . . . . . . . . . . . . 121
6.11.1 Inter Channel Interference . . . . . . . . . . . . . . . . . . . . 121
6.11.2 Power Ampli er Nonlinearity . . . . . . . . . . . . . . . . . . 122
6.12 Receiver performance in multipath channels . . . . . . . . . . . . . . 122
6.12.1 Bit Error Rate and Symbol Error Rate . . . . . . . . . . . . . 123
6.13 Example of a Multicarrier Modulation: OFDM . . . . . . . . . . . . 123
6.13.1 Orthogonality of Signals . . . . . . . . . . . . . . . . . . . . . 125
6.13.2 Mathematical Description of OFDM . . . . . . . . . . . . . . 125
6.14 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 127
6.15 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 128
7 Techniques to Mitigate Fading E ects 129
7.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 129
7.2 Equalization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 130
7.2.1 A Mathematical Framework . . . . . . . . . . . . . . . . . . . 131
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7.2.2 Zero Forcing Equalization . . . . . . . . . . . . . . . . . . . . 132
7.2.3 A Generic Adaptive Equalizer . . . . . . . . . . . . . . . . . . 132
7.2.4 Choice of Algorithms for Adaptive Equalization . . . . . . . . 134
7.3 Diversity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 136
7.3.1 Di erent Types of Diversity . . . . . . . . . . . . . . . . . . . 137
7.4 Channel Coding . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 143
7.4.1 Shannon's Channel Capacity Theorem . . . . . . . . . . . . . 143
7.4.2 Block Codes . . . . . . . . . . . . . . . . . . . . . . . . . . . . 144
7.4.3 Convolutional Codes . . . . . . . . . . . . . . . . . . . . . . . 152
7.4.4 Concatenated Codes . . . . . . . . . . . . . . . . . . . . . . . 155
7.5 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 156
7.6 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 156
8 Multiple Access Techniques 157
8.1 Multiple Access Techniques for Wireless Communication . . . . . . . 157
8.1.1 Narrowband Systems . . . . . . . . . . . . . . . . . . . . . . . 158
8.1.2 Wideband Systems . . . . . . . . . . . . . . . . . . . . . . . . 158
8.2 Frequency Division Multiple Access . . . . . . . . . . . . . . . . . . . 159
8.2.1 FDMA/FDD in AMPS . . . . . . . . . . . . . . . . . . . . . 160
8.2.2 FDMA/TDD in CT2 . . . . . . . . . . . . . . . . . . . . . . . 160
8.2.3 FDMA and Near-Far Problem . . . . . . . . . . . . . . . . . 160
8.3 Time Division Multiple Access . . . . . . . . . . . . . . . . . . . . . 161
8.3.1 TDMA/FDD in GSM . . . . . . . . . . . . . . . . . . . . . . 161
8.3.2 TDMA/TDD in DECT . . . . . . . . . . . . . . . . . . . . . 162
8.4 Spread Spectrum Multiple Access . . . . . . . . . . . . . . . . . . . . 163
8.4.1 Frequency Hopped Multiple Access (FHMA) . . . . . . . . . 163
8.4.2 Code Division Multiple Access . . . . . . . . . . . . . . . . . 163
8.4.3 CDMA and Self-interference Problem . . . . . . . . . . . . . 164
8.4.4 CDMA and Near-Far Problem . . . . . . . . . . . . . . . . . 165
8.4.5 Hybrid Spread Spectrum Techniques . . . . . . . . . . . . . . 165
8.5 Space Division Multiple Access . . . . . . . . . . . . . . . . . . . . . 166
8.6 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 166
8.7 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 167
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List of Figures
1.1 The worldwide mobile subscriber chart. . . . . . . . . . . . . . . . . 2
1.2 Basic mobile communication structure. . . . . . . . . . . . . . . . . . 3
1.3 The basic radio transmission techniques: (a) simplex, (b) half duplex
and (c) full duplex. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
1.4 (a) Frequency division duplexing and (b) time division duplexing. . . 6
1.5 Basic Cellular Structure. . . . . . . . . . . . . . . . . . . . . . . . . . 7
2.1 Data transmission with Bluetooth. . . . . . . . . . . . . . . . . . . . 20
3.1 Footprint of cells showing the overlaps and gaps. . . . . . . . . . . . 24
3.2 Frequency reuse technique of a cellular system. . . . . . . . . . . . . 25
3.3 Hando scenario at two adjacent cell boundary. . . . . . . . . . . . . 29
3.4 Hando process associated with power levels, when the user is going
from i-th cell to j-th cell. . . . . . . . . . . . . . . . . . . . . . . . . . 30
3.5 Hando process with a rectangular cell inclined at an angle . . . . . 31
3.6 First tier of co-channel interfering cells . . . . . . . . . . . . . . . . . 37
3.7 Splitting of congested seven-cell clusters. . . . . . . . . . . . . . . . . 41
3.8 A cell divided into three 120o sectors. . . . . . . . . . . . . . . . . . 43
3.9 A seven-cell cluster with 60o sectors. . . . . . . . . . . . . . . . . . . 44
3.10 The micro-cell zone concept. . . . . . . . . . . . . . . . . . . . . . . . 47
3.11 The bu erless J-channel trunked radio system. . . . . . . . . . . . . 49
3.12 Discrete-time Markov chain for the M/M/J/J trunked radio system. 49
4.1 Free space propagation model, showing the near and far elds. . . . 55
4.2 Two-ray re
ection model. . . . . . . . . . . . . . . . . . . . . . . . . 59
4.3 Phasor diagram of electric elds. . . . . . . . . . . . . . . . . . . . . 61
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4.4 Equivalent phasor diagram of Figure 4.3. . . . . . . . . . . . . . . . . 61
4.5 Huygen's secondary wavelets. . . . . . . . . . . . . . . . . . . . . . . 64
4.6 Di raction through a sharp edge. . . . . . . . . . . . . . . . . . . . . 65
4.7 Fresnel zones. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 66
4.8 Knife-edge Di raction Model . . . . . . . . . . . . . . . . . . . . . . 68
5.1 Illustration of Doppler e ect. . . . . . . . . . . . . . . . . . . . . . . 79
5.2 A generic transmitted pulsed RF signal. . . . . . . . . . . . . . . . . 83
5.3 Relationship among di erent channel functions. . . . . . . . . . . . . 85
5.4 Direct RF pulsed channel IR measurement. . . . . . . . . . . . . . . 86
5.5 Frequency domain channel IR measurement. . . . . . . . . . . . . . . 87
5.6 Two ray NLoS multipath, resulting in Rayleigh fading. . . . . . . . . 91
5.7 Rayleigh probability density function. . . . . . . . . . . . . . . . . . 93
5.8 Ricean probability density function. . . . . . . . . . . . . . . . . . . 93
5.9 Nakagami probability density function. . . . . . . . . . . . . . . . . . 94
5.10 Schematic representation of level crossing with a Rayleigh fading envelope
at 10 Hz Doppler spread. . . . . . . . . . . . . . . . . . . . . 95
5.11 Clarke and Gan's model for Rayleigh fading generation using quadrature
amplitude modulation with (a) RF Doppler lter, and, (b) baseband
Doppler lter. . . . . . . . . . . . . . . . . . . . . . . . . . . . 97
5.12 Rayleigh fading model to get both the
at and frequency selective
channel conditions. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 98
5.13 Two-ray Rayleigh fading model. . . . . . . . . . . . . . . . . . . . . . 99
6.1 BPSK signal constellation. . . . . . . . . . . . . . . . . . . . . . . . . 107
6.2 QPSK signal constellation. . . . . . . . . . . . . . . . . . . . . . . . . 108
6.3 QPSK transmitter. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 108
6.4 DQPSK constellation diagram. . . . . . . . . . . . . . . . . . . . . . 109
6.5 Scematic of the line coding techniques. . . . . . . . . . . . . . . . . . 111
6.6 Rectangular Pulse . . . . . . . . . . . . . . . . . . . . . . . . . . . . 112
6.7 Raised Cosine Pulse. . . . . . . . . . . . . . . . . . . . . . . . . . . . 113
6.8 Phase tree of 1101000 CPFSK sequence. . . . . . . . . . . . . . . . . 118
6.9 Spectrum of MSK . . . . . . . . . . . . . . . . . . . . . . . . . . . . 118
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6.10 GMSK generation scheme. . . . . . . . . . . . . . . . . . . . . . . . . 119
6.11 A simple GMSK receiver. . . . . . . . . . . . . . . . . . . . . . . . . 120
6.12 Spectrum of GMSK scheme. . . . . . . . . . . . . . . . . . . . . . . . 121
6.13 OFDM Transmitter and Receiver Block Diagram. . . . . . . . . . . . 127
7.1 A general framework of fading e ects and their mitigation techniques. 130
7.2 A generic adaptive equalizer. . . . . . . . . . . . . . . . . . . . . . . 133
7.3 Receiver selection diversity, with M receivers. . . . . . . . . . . . . . 137
7.4 Maximal ratio combining technique. . . . . . . . . . . . . . . . . . . 140
7.5 RAKE receiver. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 142
7.6 A convolutional encoder with n=2 and k=1. . . . . . . . . . . . . . . 153
7.7 State diagram representation of a convolutional encoder. . . . . . . . 153
7.8 Tree diagram representation of a convolutional encoder. . . . . . . . 154
7.9 Trellis diagram of a convolutional encoder. . . . . . . . . . . . . . . . 154
7.10 Block diagram of a turbo encoder. . . . . . . . . . . . . . . . . . . . 155
8.1 The basic concept of FDMA. . . . . . . . . . . . . . . . . . . . . . . 159
8.2 The basic concept of TDMA. . . . . . . . . . . . . . . . . . . . . . . 162
8.3 The basic concept of CDMA. . . . . . . . . . . . . . . . . . . . . . . 164
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List of Tables
2.1 Main WCDMA parameters . . . . . . . . . . . . . . . . . . . . . . . 16
7.1 Finite eld elements for US-CDPD . . . . . . . . . . . . . . . . . . . 152
8.1 MA techniques in di erent wireless communication systems . . . . . 158
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Chapter 1
Introductory Concepts
1.1 Introduction
Communication is one of the integral parts of science that has always been a focus
point for exchanging information among parties at locations physically apart. After
its discovery, telephones have replaced the telegrams and letters. Similarly, the term
`mobile' has completely revolutionized the communication by opening up innovative
applications that are limited to one's imagination. Today, mobile communication
has become the backbone of the society. All the mobile system technologies have
improved the way of living. Its main plus point is that it has privileged a common
mass of society. In this chapter, the evolution as well as the fundamental techniques
of the mobile communication is discussed.
1.2 Evolution of Mobile Radio Communications
The rst wireline telephone system was introduced in the year 1877. Mobile communication
systems as early as 1934 were based on Amplitude Modulation (AM)
schemes and only certain public organizations maintained such systems. With the
demand for newer and better mobile radio communication systems during the World
War II and the development of Frequency Modulation (FM) technique by Edwin
Armstrong, the mobile radio communication systems began to witness many new
changes. Mobile telephone was introduced in the year 1946. However, during its
initial three and a half decades it found very less market penetration owing to high
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Figure 1.1: The worldwide mobile subscriber chart.
costs and numerous technological drawbacks. But with the development of the cellular
concept in the 1960s at the Bell Laboratories, mobile communications began to
be a promising eld of expanse which could serve wider populations. Initially, mobile
communication was restricted to certain o cial users and the cellular concept was
never even dreamt of being made commercially available. Moreover, even the growth
in the cellular networks was very slow. However, with the development of newer and
better technologies starting from the 1970s and with the mobile users now connected
to the Public Switched Telephone Network (PSTN), there has been an astronomical
growth in the cellular radio and the personal communication systems. Advanced
Mobile Phone System (AMPS) was the rst U.S. cellular telephone system and it
was deployed in 1983. Wireless services have since then been experiencing a 50%
per year growth rate. The number of cellular telephone users grew from 25000 in
1984 to around 3 billion in the year 2007 and the demand rate is increasing day by
day. A schematic of the subscribers is shown in Fig. 1.1.
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Figure 1.2: Basic mobile communication structure.
1.3 Present Day Mobile Communication
Since the time of wireless telegraphy, radio communication has been used extensively.
Our society has been looking for acquiring mobility in communication since then.
Initially the mobile communication was limited between one pair of users on single
channel pair. The range of mobility was de ned by the transmitter power, type of
antenna used and the frequency of operation. With the increase in the number of
users, accommodating them within the limited available frequency spectrum became
a major problem. To resolve this problem, the concept of cellular communication
was evolved. The present day cellular communication uses a basic unit called cell.
Each cell consists of small hexagonal area with a base station located at the center
of the cell which communicates with the user. To accommodate multiple users
Time Division multiple Access (TDMA), Code Division Multiple Access (CDMA),
Frequency Division Multiple Access (FDMA) and their hybrids are used. Numerous
mobile radio standards have been deployed at various places such as AMPS, PACS,
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Figure 1.3: The basic radio transmission techniques: (a) simplex, (b) half duplex
and (c) full duplex.
GSM, NTT, PHS and IS-95, each utilizing di erent set of frequencies and allocating
di erent number of users and channels.
1.4 Fundamental Techniques
By de nition, mobile radio terminal means any radio terminal that could be moved
during its operation. Depending on the radio channel, there can be three di erent
types of mobile communication. In general, however, a Mobile Station (MS)
or subscriber unit communicates to a xed Base Station (BS) which in turn communicates
to the desired user at the other end. The MS consists of transceiver,
control circuitry, duplexer and an antenna while the BS consists of transceiver and
channel multiplexer along with antennas mounted on the tower. The BS are also
linked to a power source for the transmission of the radio signals for communication
and are connected to a xed backbone network. Figure 1.2 shows a basic mobile
communication with low power transmitters/receivers at the BS, the MS and also
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the Mobile Switching Center (MSC). The MSC is sometimes also called Mobile Telephone
Switching O ce (MTSO). The radio signals emitted by the BS decay as the
signals travel away from it. A minimum amount of signal strength is needed in
order to be detected by the mobile stations or mobile sets which are the hand-held
personal units (portables) or those installed in the vehicles (mobiles). The region
over which the signal strength lies above such a threshold value is known as the
coverage area of a BS. The xed backbone network is a wired network that links all
the base stations and also the landline and other telephone networks through wires.
1.4.1 Radio Transmission Techniques
Based on the type of channels being utilized, mobile radio transmission systems may
be classi ed as the following three categories which is also shown in Fig. 1.3:
Simplex System: Simplex systems utilize simplex channels i.e., the communication
is unidirectional. The rst user can communicate with the second
user. However, the second user cannot communicate with the rst user. One
example of such a system is a pager.
Half Duplex System: Half duplex radio systems that use half duplex radio
channels allow for non-simultaneous bidirectional communication. The rst
user can communicate with the second user but the second user can communicate
to the rst user only after the rst user has nished his conversation.
At a time, the user can only transmit or receive information. A walkie-talkie
is an example of a half duplex system which uses `push to talk' and `release to
listen' type of switches.
Full Duplex System: Full duplex systems allow two way simultaneous communications.
Both the users can communicate to each other simultaneously.
This can be done by providing two simultaneous but separate channels to both
the users. This is possible by one of the two following methods.
{ Frequency Division Duplexing (FDD): FDD supports two-way radio
communication by using two distinct radio channels. One frequency channel
is transmitted downstream from the BS to the MS (forward channel).
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Figure 1.4: (a) Frequency division duplexing and (b) time division duplexing.
A second frequency is used in the upstream direction and supports transmission
from the MS to the BS (reverse channel). Because of the pairing of
frequencies, simultaneous transmission in both directions is possible. To
mitigate self-interference between upstream and downstream transmissions,
a minimum amount of frequency separation must be maintained
between the frequency pair, as shown in Fig. 1.4.
{ Time Division Duplexing (TDD): TDD uses a single frequency band
to transmit signals in both the downstream and upstream directions.
TDD operates by toggling transmission directions over a time interval.
This toggling takes place very rapidly and is imperceptible to the user.
A full duplex mobile system can further be subdivided into two category: a
single MS for a dedicated BS, and many MS for a single BS. Cordless telephone
systems are full duplex communication systems that use radio to connect to a
portable handset to a single dedicated BS, which is then connected to a dedicated
telephone line with a speci c telephone number on the Public Switched
Telephone Network (PSTN). A mobile system, in general, on the other hand,
is the example of the second category of a full duplex mobile system where
many users connect among themselves via a single BS.
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Figure 1.5: Basic Cellular Structure.
1.5 How a Mobile Call is Actually Made?
In order to know how a mobile call is made, we should rst look into the basics of
cellular concept and main operational channels involved in making a call. These are
given below.
1.5.1 Cellular Concept
Cellular telephone systems must accommodate a large number of users over a large
geographic area with limited frequency spectrum, i.e., with limited number of channels.
If a single transmitter/ receiver is used with only a single base station, then
su cient amount of power may not be present at a huge distance from the BS.
For a large geographic coverage area, a high powered transmitter therefore has to
be used. But a high power radio transmitter causes harm to environment. Mobile
communication thus calls for replacing the high power transmitters by low power
transmitters by dividing the coverage area into small segments, called cells. Each
cell uses a certain number of the available channels and a group of adjacent cells
together use all the available channels. Such a group is called a cluster. This cluster
can repeat itself and hence the same set of channels can be used again and again.
Each cell has a low power transmitter with a coverage area equal to the area of the
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cell. This technique of substituting a single high powered transmitter by several low
powered transmitters to support many users is the backbone of the cellular concept.
1.5.2 Operational Channels
In each cell, there are four types of channels that take active part during a mobile
call. These are:
Forward Voice Channel (FVC): This channel is used for the voice transmission
from the BS to the MS.
Reverse Voice Channel (RVC): This is used for the voice transmission
from the MS to the BS.
Forward Control Channel (FCC): Control channels are generally used
for controlling the activity of the call, i.e., they are used for setting up calls
and to divert the call to unused voice channels. Hence these are also called
setup channels. These channels transmit and receive call initiation and service
request messages. The FCC is used for control signaling purpose from the BS
to MS.
Reverse Control Channel (RCC): This is used for the call control purpose
from the MS to the BS. Control channels are usually monitored by mobiles.
1.5.3 Making a Call
When a mobile is idle, i.e., it is not experiencing the process of a call, then it searches
all the FCCs to determine the one with the highest signal strength. The mobile
then monitors this particular FCC. However, when the signal strength falls below
a particular threshold that is insu cient for a call to take place, the mobile again
searches all the FCCs for the one with the highest signal strength. For a particular
country or continent, the control channels will be the same. So all mobiles in that
country or continent will search among the same set of control channels. However,
when a mobile moves to a di erent country or continent, then the control channels
for that particular location will be di erent and hence the mobile will not work.
Each mobile has a mobile identi cation number (MIN). When a user wants to
make a call, he sends a call request to the MSC on the reverse control channel. He
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also sends the MIN of the person to whom the call has to be made. The MSC then
sends this MIN to all the base stations. The base station transmits this MIN and all
the mobiles within the coverage area of that base station receive the MIN and match
it with their own. If the MIN matches with a particular MS, that mobile sends an
acknowledgment to the BS. The BS then informs the MSC that the mobile is within
its coverage area. The MSC then instructs the base station to access speci c unused
voice channel pair. The base station then sends a message to the mobile to move to
the particular channels and it also sends a signal to the mobile for ringing.
In order to maintain the quality of the call, the MSC adjusts the transmitted
power of the mobile which is usually expressed in dB or dBm. When a mobile moves
from the coverage area of one base station to the coverage area of another base station
i.e., from one cell to another cell, then the signal strength of the initial base
station may not be su cient to continue the call in progress. So the call has to be
transferred to the other base station. This is called hando . In such cases, in order
to maintain the call, the MSC transfers the call to one of the unused voice channels
of the new base station or it transfers the control of the current voice channels to
the new base station.
Ex. 1: Suppose a mobile unit transmits 10 W power at a certain place. Express this
power in terms of dBm.
Solution: Usually, 1 mW power developed over a 100
 load is equivalently called
0 dBm power. 1 W is equivalent to 0 dB, i.e., 10 log10(1W) = 0dB. Thus,
1W = 103mW = 30dBm = 0dB. This means, xdB = (x + 30)dBm. Hence,
10W = 10 log10(10W) = 10dB = 40dBm.
Ex. 2: Among a pager, a cordless phone and a mobile phone, which device would
have the (i) shortest, and, (ii) longest battery life? Justify.
Solution: The `pager' would have the longest and the `mobile phone' would have the
shortest battery life. (justi cation is left on the readers)
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1.6 Future Trends
Tremendous changes are occurring in the area of mobile radio communications, so
much so that the mobile phone of yesterday is rapidly turning into a sophisticated
mobile device capable of more applications than PCs were capable of only a few
years ago. Rapid development of the Internet with its new services and applications
has created fresh challenges for the further development of mobile communication
systems. Further enhancements in modulation schemes will soon increase the Internet
access rates on the mobile from current 1.8 Mbps to greater than 10 Mbps.
Bluetooth is rapidly becoming a common feature in mobiles for local connections.
The mobile communication has provided global connectivity to the people at
a lower cost due to advances in the technology and also because of the growing
competition among the service providers. We would review certain major features
as well as standards of the mobile communication till the present day technology in
the next chapter.
1.7 References
1. T. S. Rappaport, Wireless Communications: Principles and Practice, 2nd ed.
Singapore: Pearson Education, Inc., 2002.
2. K. Feher, Wireless Digital Communications: Modulation and Spread Spectrum
Applications. Upper Saddle River, NJ: Prentice Hall, 1995.
3. J. G. Proakis, Digital Communications, 4th ed. NY: McGraw Hill, 2000.
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Chapter 2
Modern Wireless
Communication Systems
At the initial phase, mobile communication was restricted to certain o cial users and
the cellular concept was never even dreamt of being made commercially available.
Moreover, even the growth in the cellular networks was very slow. However, with
the development of newer and better technologies starting from the 1970s and with
the mobile users now connected to the PSTN, there has been a remarkable growth
in the cellular radio. However, the spread of mobile communication was very fast
in the 1990s when the government throughout the world provided radio spectrum
licenses for Personal Communication Service (PCS) in 1.8 - 2 GHz frequency band.
2.1 1G: First Generation Networks
The rst mobile phone system in the market was AMPS. It was the rst U.S. cellular
telephone system, deployed in Chicago in 1983. The main technology of this rst
generation mobile system was FDMA/FDD and analog FM.
2.2 2G: Second Generation Networks
Digital modulation formats were introduced in this generation with the main technology
as TDMA/FDD and CDMA/FDD. The 2G systems introduced three popular
TDMA standards and one popular CDMA standard in the market. These are as
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follows:
2.2.1 TDMA/FDD Standards
(a) Global System for Mobile (GSM): The GSM standard, introduced by Groupe
Special Mobile, was aimed at designing a uniform pan-European mobile system. It
was the rst fully digital system utilizing the 900 MHz frequency band. The initial
GSM had 200 KHz radio channels, 8 full-rate or 16 half-rate TDMA channels per
carrier, encryption of speech, low speed data services and support for SMS for which
it gained quick popularity.
(b) Interim Standard 136 (IS-136): It was popularly known as North American
Digital Cellular (NADC) system. In this system, there were 3 full-rate TDMA users
over each 30 KHz channel. The need of this system was mainly to increase the
capacity over the earlier analog (AMPS) system.
(c) Paci c Digital Cellular (PDC): This standard was developed as the counterpart
of NADC in Japan. The main advantage of this standard was its low transmission
bit rate which led to its better spectrum utilization.
2.2.2 CDMA/FDD Standard
Interim Standard 95 (IS-95): The IS-95 standard, also popularly known as CDMAOne,
uses 64 orthogonally coded users and codewords are transmitted simultaneously
on each of 1.25 MHz channels. Certain services that have been standardized as a
part of IS-95 standard are: short messaging service, slotted paging, over-the-air
activation (meaning the mobile can be activated by the service provider without
any third party intervention), enhanced mobile station identities etc.
2.2.3 2.5G Mobile Networks
In an e ort to retro t the 2G standards for compatibility with increased throughput
rates to support modern Internet application, the new data centric standards were
developed to be overlaid on 2G standards and this is known as 2.5G standard.
Here, the main upgradation techniques are:
supporting higher data rate transmission for web browsing
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supporting e-mail tra c
enabling location-based mobile service
2.5G networks also brought into the market some popular application, a few of
which are: Wireless Application Protocol (WAP), General Packet Radio Service
(GPRS), High Speed Circuit Switched Dada (HSCSD), Enhanced Data rates for
GSM Evolution (EDGE) etc.
2.3 3G: Third Generation Networks
3G is the third generation of mobile phone standards and technology, superseding
2.5G. It is based on the International Telecommunication Union (ITU) family
of standards under the International Mobile Telecommunications-2000 (IMT-2000).
ITU launched IMT-2000 program, which, together with the main industry and standardization
bodies worldwide, targets to implement a global frequency band that
would support a single, ubiquitous wireless communication standard for all countries,
to provide the framework for the de nition of the 3G mobile systems.Several
radio access technologies have been accepted by ITU as part of the IMT-2000 framework.
3G networks enable network operators to o er users a wider range of more advanced
services while achieving greater network capacity through improved spectral
e ciency. Services include wide-area wireless voice telephony, video calls, and broadband
wireless data, all in a mobile environment. Additional features also include
HSPA data transmission capabilities able to deliver speeds up to 14.4Mbit/s on the
down link and 5.8Mbit/s on the uplink.
3G networks are wide area cellular telephone networks which evolved to incorporate
high-speed internet access and video telephony. IMT-2000 de nes a set of
technical requirements for the realization of such targets, which can be summarized
as follows:
high data rates: 144 kbps in all environments and 2 Mbps in low-mobility and
indoor environments
symmetrical and asymmetrical data transmission
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circuit-switched and packet-switched-based services
speech quality comparable to wire-line quality
improved spectral e ciency
several simultaneous services to end users for multimedia services
seamless incorporation of second-generation cellular systems
global roaming
open architecture for the rapid introduction of new services and technology.
2.3.1 3G Standards and Access Technologies
As mentioned before, there are several di erent radio access technologies de ned
within ITU, based on either CDMA or TDMA technology. An organization called
3rd Generation Partnership Project (3GPP) has continued that work by de ning a
mobile system that ful lls the IMT-2000 standard. This system is called Universal
Mobile Telecommunications System (UMTS). After trying to establish a single 3G
standard, ITU nally approved a family of ve 3G standards, which are part of the
3G framework known as IMT-2000:
W-CDMA
CDMA2000
TD-SCDMA
Europe, Japan, and Asia have agreed upon a 3G standard called the Universal
Mobile Telecommunications System (UMTS), which is WCDMA operating at 2.1
GHz. UMTS and WCDMA are often used as synonyms. In the USA and other
parts of America, WCDMA will have to use another part of the radio spectrum.
2.3.2 3G W-CDMA (UMTS)
WCDMA is based on DS-CDMA (direct sequencecode division multiple access) technology
in which user-information bits are spread over a wide bandwidth (much
larger than the information signal bandwidth) by multiplying the user data with
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the spreading code. The chip (symbol rate) rate of the spreading sequence is 3.84
Mcps, which, in the WCDMA system deployment is used together with the 5-MHz
carrier spacing. The processing gain term refers to the relationship between the
signal bandwidth and the information bandwidth. Thus, the name wideband is
derived to di erentiate it from the 2G CDMA (IS-95), which has a chip rate of
1.2288 Mcps. In a CDMA system, all users are active at the same time on the same
frequency and are separated from each other with the use of user speci c spreading
codes.
The wide carrier bandwidth of WCDMA allows supporting high user-data rates
and also has certain performance bene ts, such as increased multipath diversity.
The actual carrier spacing to be used by the operator may vary on a 200-kHz grid
between approximately 4.4 and 5 MHz, depending on spectrum arrangement and
the interference situation.
In WCDMA each user is allocated frames of 10 ms duration, during which the
user-data rate is kept constant. However, the data rate among the users can change
from frame to frame. This fast radio capacity allocation (or the limits for variation in
the uplink) is controlled and coordinated by the radio resource management (RRM)
functions in the network to achieve optimum throughput for packet data services
and to ensure su cient quality of service (QoS) for circuit-switched users. WCDMA
supports two basic modes of operation: FDD and TDD. In the FDD mode, separate
5-MHz carrier frequencies with duplex spacing are used for the uplink and downlink,
respectively, whereas in TDD only one 5-MHz carrier is time shared between the uplink
and the downlink. WCDMA uses coherent detection based on the pilot symbols
and/or common pilot. WCDMA allows many performance- enhancement methods
to be used, such as transmit diversity or advanced CDMA receiver concepts.Table
summaries the main WCDMA parameters.
The support for handovers (HO) between GSM and WCDMA is part of the rst
standard version. This means that all multi-mode WCDMA/GSM terminals will
support measurements from the one system while camped on the other one. This
allows networks using both WCDMA and GSM to balance the load between the
networks and base the HO on actual measurements from the terminals for di erent
radio conditions in addition to other criteria available.
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Table 2.1: Main WCDMA parameters
Multiple access method DS-CDMA
Duplexing method Frequency division duplex/time division
duplex
Base station synchronisation Asynchronous operation
Chip rate 3.84 Mcps
Frame length 10 ms
Service multiplexing Multiple services with di erent quality of
service requirements multiplexed on one
connection
Multi-rate concept Variable spreading factor and multicode
Detection Coherent using pilot symbols or common
pilot
Multi-user detection, smart antennas Supported by the standard, optional in the
implementation
The world's rst commercial W-CDMA service, FoMA, was launched by NTT
DoCoMo in Japan in 2001. FoMA is the short name for Freedom of Mobile Multimedia
Access, is the brand name for the 3G services being o ered by Japanese
mobile phone operator NTT DoCoMo. Elsewhere, W-CDMA deployments have
been exclusively UMTS based.
UMTS or W-CDMA, assures backward compatibility with the second generation
GSM, IS-136 and PDC TDMA technologies, as well as all 2.5G TDMA technologies.
The network structure and bit level packaging of GSM data is retained by W-CDMA,
with additional capacity and bandwidth provided by a new CDMA air interface.
2.3.3 3G CDMA2000
Code division multiple access 2000 is the natural evolution of IS-95 (cdmaOne). It
includes additional functionality that increases its spectral e ciency and data rate
capability.(code division multiple access) is a mobile digital radio technology where
channels are de ned with codes (PN sequences). CDMA permits many simultaneous
transmitters on the same frequency channel. Since more phones can be served by
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fewer cell sites, CDMA-based standards have a signi cant economic advantage over
TDMA- or FDMA-based standards. This standard is being developed by Telecommunications
Industry Association (TIA) of US and is is standardized by 3GPP2.
The main CDMA2000 standards are: CDMA2000 1xRTT,CDMA2000 1xEV and
CDMA2000 EV-DV. These are the approved radio interfaces for the ITU's IMT-2000
standard. In the following, a brief discussion about all these standards is given.
CDMA2000 1xRTT: RTT stands for Radio Transmission Technology and the
designation "1x", meaning "1 times Radio Transmission Technology", indicates the
same RF bandwidth as IS-95.The main features of CDMA2000 1X are as follows:
Supports an instantaneous data rate upto 307kpbs for a user in packet mode
and a typical throughput rates of 144kbps per user,depending on the number
of user, the velociy of user and the propagating conditions.
Supports up to twice as many voice users a the 2G CDMA standard
Provides the subscriber unit with upto two times the standby time for longer
lasting battery life.
CDMA2000 EV: This is an evolutionary advancement of CDMA with the
following characteristics:
Provides CDMA carriers with the option of installing radio channels with data
only (CDMA2000 EV-DO) and with data and voice (CDMA2000 EV-DV) .
The cdma2000 1xEV-DO supports greater than 2.4Mbps of instantaneous
high-speed packet throughput per user on a CDMA channel, although the
user data rates are much lower and highly dependent on other factors.
CDMA2000 EV-DV can o er data rates upto 144kbps with about twice as
many voice channels as IS-95B.
CDMA2000 3x is (also known as EV-DO Rev B) is a multi-carrier evolution.
It has higher rates per carrier (up to 4.9 Mbit/s on the downlink per carrier).
Typical deployments are expected to include 3 carriers for a peak rate of 14.7
Mbit/s.Higher rates are possible by bundling multiple channels together. It
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enhances the user experience and enables new services such as high de nition
video streaming.
Uses statistical multiplexing across channels to further reduce latency, enhancing
the experience for latency-sensitive services such as gaming, video
telephony, remote console sessions and web browsing.
It provides increased talk-time and standby time.
The interference from the adjacent sectors is reduced by hybrid frequency reuse
and improves the rates that can be o ered, especially to users at the edge
of the cell.
It has e cient support for services that have asymmetric download and upload
requirements (i.e. di erent data rates required in each direction) such as le
transfers, web browsing, and broadband multimedia content delivery.
2.3.4 3G TD-SCDMA
Time Division-Synchronous Code Division Multiple Access, or TD-SCDMA, is a
3G mobile telecommunications standard, being pursued in the People's Republic of
China by the Chinese Academy of Telecommunications Technology (CATT). This
proposal was adopted by ITU as one of the 3G options in late 1999. TD-SCDMA is
based on spread spectrum technology.
TD-SCDMA uses TDD, in contrast to the FDD scheme used by W-CDMA.
By dynamically adjusting the number of timeslots used for downlink and uplink,
the system can more easily accommodate asymmetric tra c with di erent data
rate requirements on downlink and uplink than FDD schemes. Since it does not
require paired spectrum for downlink and uplink, spectrum allocation
exibility is
also increased. Also, using the same carrier frequency for uplink and downlink means
that the channel condition is the same on both directions, and the base station can
deduce the downlink channel information from uplink channel estimates, which is
helpful to the application of beamforming techniques.
TD-SCDMA also uses TDMA in addition to the CDMA used in WCDMA. This
reduces the number of users in each timeslot, which reduces the implementation
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complexity of multiuser detection and beamforming schemes, but the non-continuous
transmission also reduces coverage (because of the higher peak power needed), mobility
(because of lower power control frequency) and complicates radio resource
management algorithms.
The "S" in TD-SCDMA stands for "synchronous", which means that uplink signals
are synchronized at the base station receiver, achieved by continuous timing
adjustments. This reduces the interference between users of the same timeslot using
di erent codes by improving the orthogonality between the codes, therefore increasing
system capacity, at the cost of some hardware complexity in achieving uplink
synchronization.
2.4 Wireless Transmission Protocols
There are several transmission protocols in wireless manner to achieve di erent
application oriented tasks. Below, some of these applications are given.
2.4.1 Wireless Local Loop (WLL) and LMDS
Microwave wireless links can be used to create a wireless local loop. The local loop
can be thought of as the "last mile" of the telecommunication network that resides
between the central o ce (CO) and the individual homes and business in close
proximity to the CO. An advantage of WLL technology is that once the wireless
equipment is paid for, there are no additional costs for transport between the CO
and the customer premises equipment. Many new services have been proposed and
this includes the concept of Local Multipoint Distribution Service (LMDS), which
provides broadband telecommunication access in the local exchange.
2.4.2 Bluetooth
Facilitates ad-hoc data transmission over short distances from xed and mobile
devices as shown in Figure 2.1
Uses a radio technology called frequency hopping spread spectrum. It chops up
the data being sent and transmits chunks of it on up to 79 di erent frequencies.
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Figure 2.1: Data transmission with Bluetooth.
In its basic mode, the modulation is Gaussian frequency shift keying (GFSK).
It can achieve a gross data rate of 1 Mb/s
Primarily designed for low power consumption, with a short range (powerclass-
dependent: 1 meter, 10 meters, 100 meters) based on low-cost transceiver
microchips in each device
2.4.3 Wireless Local Area Networks (W-LAN)
IEEE 802.11 WLAN uses ISM band (5.275-5.825GHz)
Uses 11Mcps DS-SS spreading and 2Mbps user data rates (will fallback to
1Mbps in noisy conditions)
IEEE 802.11a stndard provides upto 54Mbps throughput in the 5GHz band.
The DS-SS IEEE 802.11b has been called Wi-Fi. Wi-Fi networks have limited
range. A typical Wi-Fi home router using 802.11b or 802.11g with a stock
antenna might have a range of 32 m (120 ft) indoors and 95 m (300 ft) outdoors.
Range also varies with frequency band.
IEEE 802.11g uses Complementary Code Keying Orthogonal Frequency Division
Multiplexing (CCK-OFDM) standards in both 2.4GHz and 5GHz bands.
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2.4.4 WiMax
Provides upto 70 Mb/sec symmetric broadband speed without the need for
cables. The technology is based on the IEEE 802.16 standard (also called
WirelessMAN)
WiMAX can provide broadband wireless access (BWA) up to 30 miles (50 km)
for xed stations, and 3 - 10 miles (5 - 15 km) for mobile stations. In contrast,
the WiFi/802.11 wireless local area network standard is limited in most cases
to only 100 - 300 feet (30 - 100m)
The 802.16 speci cation applies across a wide range of the RF spectrum, and
WiMAX could function on any frequency below 66 GHz (higher frequencies
would decrease the range of a Base Station to a few hundred meters in an
urban environment).
2.4.5 Zigbee
ZigBee is the speci cation for a suite of high level communication protocols using
small, low-power digital radios based on the IEEE 802.15.4-2006 standard
for wireless personal area networks (WPANs), such as wireless headphones
connecting with cell phones via short-range radio.
This technology is intended to be simpler and cheaper. ZigBee is targeted at
radio-frequency (RF) applications that require a low data rate, long battery
life, and secure networking.
ZigBee operates in the industrial, scienti c and medical (ISM) radio bands;
868 MHz in Europe, 915 MHz in countries such as USA and Australia, and
2.4 GHz in most worldwide.
2.4.6 Wibree
Wibree is a digital radio technology (intended to become an open standard of
wireless communications) designed for ultra low power consumption (button
cell batteries) within a short range (10 meters / 30 ft) based around low-cost
transceiver microchips in each device.
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Wibree is known as Bluetooth with low energy technology.
It operates in 2.4 GHz ISM band with physical layer bit rate of 1 Mbps.
2.5 Conclusion: Beyond 3G Networks
Beyond 3G networks, or 4G (Fourth Generation), represent the next complete evolution
in wireless communications. A 4G system will be able to provide a comprehensive
IP solution where voice, data and streamed multimedia can be given to users
at higher data rates than previous generations.There is no formal de nition for 4G ;
however, there are certain objectives that are projected for 4G. It will be capable of
providing between 100 Mbit/s and 1 Gbit/s speeds both indoors and outdoors, with
premium quality and high security. It would also support systems like multicarrier
communication, MIMO and UWB.
2.6 References
1. T. S. Rappaport, Wireless Communications: Principles and Practice, 2nd ed.
Singapore: Pearson Education, Inc., 2002.
2. W. C. Lee, Mobile Communications Engineering, 2nd ed. New Delhi: Tata
McGraw-Hill, 2008.
3. R. Pandya, Mobile and Personal Communication Systems and Services, 4th
ed. New Delhi: PHI, 2004.
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Chapter 3
The Cellular Engineering
Fundamentals
3.1 Introduction
In Chapter 1, we have seen that the technique of substituting a single high power
transmitter by several low power transmitters to support many users is the backbone
of the cellular concept. In practice, the following four parameters are most important
while considering the cellular issues: system capacity, quality of service, spectrum
e ciency and power management. Starting from the basic notion of a cell, we would
deal with these parameters in the context of cellular engineering in this chapter.
3.2 What is a Cell?
The power of the radio signals transmitted by the BS decay as the signals travel
away from it. A minimum amount of signal strength (let us say, x dB) is needed in
order to be detected by the MS or mobile sets which may the hand-held personal
units or those installed in the vehicles. The region over which the signal strength
lies above this threshold value x dB is known as the coverage area of a BS and
it must be a circular region, considering the BS to be isotropic radiator. Such a
circle, which gives this actual radio coverage, is called the foot print of a cell (in
reality, it is amorphous). It might so happen that either there may be an overlap
between any two such side by side circles or there might be a gap between the
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Figure 3.1: Footprint of cells showing the overlaps and gaps.
coverage areas of two adjacent circles. This is shown in Figure 3.1. Such a circular
geometry, therefore, cannot serve as a regular shape to describe cells. We need a
regular shape for cellular design over a territory which can be served by 3 regular
polygons, namely, equilateral triangle, square and regular hexagon, which can cover
the entire area without any overlap and gaps. Along with its regularity, a cell must
be designed such that it is most reliable too, i.e., it supports even the weakest mobile
with occurs at the edges of the cell. For any distance between the center and the
farthest point in the cell from it, a regular hexagon covers the maximum area. Hence
regular hexagonal geometry is used as the cells in mobile communication.
3.3 Frequency Reuse
Frequency reuse, or, frequency planning, is a technique of reusing frequencies and
channels within a communication system to improve capacity and spectral e ciency.
Frequency reuse is one of the fundamental concepts on which commercial wireless
systems are based that involve the partitioning of an RF radiating area into cells.
The increased capacity in a commercial wireless network, compared with a network
with a single transmitter, comes from the fact that the same radio frequency can be
reused in a di erent area for a completely di erent transmission.
Frequency reuse in mobile cellular systems means that frequencies allocated to
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Figure 3.2: Frequency reuse technique of a cellular system.
the service are reused in a regular pattern of cells, each covered by one base station.
The repeating regular pattern of cells is called cluster. Since each cell is designed
to use radio frequencies only within its boundaries, the same frequencies can be
reused in other cells not far away without interference, in another cluster. Such cells
are called `co-channel' cells. The reuse of frequencies enables a cellular system to
handle a huge number of calls with a limited number of channels. Figure 3.2 shows
a frequency planning with cluster size of 7, showing the co-channels cells in di erent
clusters by the same letter. The closest distance between the co-channel cells (in
di erent clusters) is determined by the choice of the cluster size and the layout of
the cell cluster. Consider a cellular system with S duplex channels available for
use and let N be the number of cells in a cluster. If each cell is allotted K duplex
channels with all being allotted unique and disjoint channel groups we have S = KN
under normal circumstances. Now, if the cluster are repeated M times within the
total area, the total number of duplex channels, or, the total number of users in the
system would be T = MS = KMN. Clearly, if K and N remain constant, then
T / M (3.1)
and, if T and K remain constant, then
N /
1
M
: (3.2)
Hence the capacity gain achieved is directly proportional to the number of times
a cluster is repeated, as shown in (3.1), as well as, for a xed cell size, small N
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decreases the size of the cluster with in turn results in the increase of the number
of clusters (3.2) and hence the capacity. However for small N, co-channel cells are
located much closer and hence more interference. The value of N is determined by
calculating the amount of interference that can be tolerated for a su cient quality
communication. Hence the smallest N having interference below the tolerated limit
is used. However, the cluster size N cannot take on any value and is given only by
the following equation
N = i2 + ij + j2; i 0; j 0; (3.3)
where i and j are integer numbers.
Ex. 1: Find the relationship between any two nearest co-channel cell distance D
and the cluster size N.
Solution: For hexagonal cells, it can be shown that the distance between two adjacent
cell centers =
p
3R, where R is the radius of any cell. The normalized co-channel
cell distance Dn can be calculated by traveling 'i' cells in one direction and then
traveling 'j' cells in anticlockwise 120o of the primary direction. Using law of vector
addition,
D2
n = j2 cos2(30o) + (i + j sin(30o))2 (3.4)
which turns out to be
Dn =
q
i2 + ij + j2 =
p
N: (3.5)
Multiplying the actual distance
p
3R between two adjacent cells with it, we get
D = Dn
p
3R =
p
3NR: (3.6)
Ex. 2: Find out the surface area of a regular hexagon with radius R, the surface
area of a large hexagon with radius D, and hence compute the total number of cells
in this large hexagon.
Hint: In general, this large hexagon with radius D encompasses the center cluster of
N cells and one-third of the cells associated with six other peripheral large hexagons.
Thus, the answer must be N + 6(N
3 ) = 3N.
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3.4 Channel Assignment Strategies
With the rapid increase in number of mobile users, the mobile service providers
had to follow strategies which ensure the e ective utilization of the limited radio
spectrum. With increased capacity and low interference being the prime objectives,
a frequency reuse scheme was helpful in achieving this objectives. A variety of
channel assignment strategies have been followed to aid these objectives. Channel
assignment strategies are classi ed into two types: xed and dynamic, as discussed
below.
3.4.1 Fixed Channel Assignment (FCA)
In xed channel assignment strategy each cell is allocated a xed number of voice
channels. Any communication within the cell can only be made with the designated
unused channels of that particular cell. Suppose if all the channels are occupied,
then the call is blocked and subscriber has to wait. This is simplest of the channel
assignment strategies as it requires very simple circuitry but provides worst channel
utilization. Later there was another approach in which the channels were borrowed
from adjacent cell if all of its own designated channels were occupied. This was
named as borrowing strategy. In such cases the MSC supervises the borrowing process
and ensures that none of the calls in progress are interrupted.
3.4.2 Dynamic Channel Assignment (DCA)
In dynamic channel assignment strategy channels are temporarily assigned for use
in cells for the duration of the call. Each time a call attempt is made from a cell the
corresponding BS requests a channel from MSC. The MSC then allocates a channel
to the requesting the BS. After the call is over the channel is returned and kept in
a central pool. To avoid co-channel interference any channel that in use in one cell
can only be reassigned simultaneously to another cell in the system if the distance
between the two cells is larger than minimum reuse distance. When compared to the
FCA, DCA has reduced the likelihood of blocking and even increased the trunking
capacity of the network as all of the channels are available to all cells, i.e., good
quality of service. But this type of assignment strategy results in heavy load on
switching center at heavy tra c condition.
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Ex. 3: A total of 33 MHz bandwidth is allocated to a FDD cellular system with
two 25 KHz simplex channels to provide full duplex voice and control channels.
Compute the number of channels available per cell if the system uses (i) 4 cell, (ii)
7 cell, and (iii) 8 cell reuse technique. Assume 1 MHz of spectrum is allocated to
control channels. Give a distribution of voice and control channels.
Solution: One duplex channel = 2 x 25 = 50 kHz of spectrum. Hence the total
available duplex channels are = 33 MHz / 50 kHz = 660 in number. Among these
channels, 1 MHz / 50 kHz = 20 channels are kept as control channels.
(a) For N = 4, total channels per cell = 660/4 = 165.
Among these, voice channels are 160 and control channels are 5 in number.
(b) For N = 7, total channels per cell are 660/7 94. Therefore, we have to go for
a more exact solution. We know that for this system, a total of 20 control channels
and a total of 640 voice channels are kept. Here, 6 cells can use 3 control channels
and the rest two can use 2 control channels each. On the other hand, 5 cells can use
92 voice channels and the rest two can use 90 voice channels each. Thus the total
solution for this case is:
6 x 3 + 1 x 2 = 20 control channels, and,
5 x 92 + 2 x 90 = 640 voice channels.
This is one solution, there might exist other solutions too.
(c) The option N = 8 is not a valid option since it cannot satisfy equation (3.3) by
two integers i and j.
3.5 Hando Process
When a user moves from one cell to the other, to keep the communication between
the user pair, the user channel has to be shifted from one BS to the other without
interrupting the call, i.e., when a MS moves into another cell, while the conversation
is still in progress, the MSC automatically transfers the call to a new FDD channel
without disturbing the conversation. This process is called as hando . A schematic
diagram of hando is given in Figure 3.3.
Processing of hando is an important task in any cellular system. Hando s
must be performed successfully and be imperceptible to the users. Once a signal
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Figure 3.3: Hando scenario at two adjacent cell boundary.
level is set as the minimum acceptable for good voice quality (Prmin), then a slightly
stronger level is chosen as the threshold (PrH)at which hando has to be made, as
shown in Figure 3.4. A parameter, called power margin, de ned as
= PrH 􀀀 Prmin (3.7)
is quite an important parameter during the hando process since this margin can
neither be too large nor too small. If is too small, then there may not be enough
time to complete the hando and the call might be lost even if the user crosses the
cell boundary.
If is too high o the other hand, then MSC has to be burdened with unnecessary
hando s. This is because MS may not intend to enter the other cell. Therefore
should be judiciously chosen to ensure imperceptible hando s and to meet other
objectives.
3.5.1 Factors In
uencing Hando s
The following factors in
uence the entire hando process:
(a) Transmitted power: as we know that the transmission power is di erent for different
cells, the hando threshold or the power margin varies from cell to cell.
(b) Received power: the received power mostly depends on the Line of Sight (LoS)
path between the user and the BS. Especially when the user is on the boundary of
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Figure 3.4: Hando process associated with power levels, when the user is going
from i-th cell to j-th cell.
the two cells, the LoS path plays a critical role in hando s and therefore the power
margin depends on the minimum received power value from cell to cell.
(c) Area and shape of the cell: Apart from the power levels, the cell structure also
a plays an important role in the hando process.
(d) Mobility of users: The number of mobile users entering or going out of a particular
cell, also xes the hando strategy of a cell.
To illustrate the reasons (c) and (d), let us consider a rectangular cell with sides R1
and R2 inclined at an angle with horizon, as shown in the Figure 3.5. Assume N1
users are having hando in horizontal direction and N2 in vertical direction per unit
length.
The number of crossings along R1 side is : (N1cos +N2sin )R1 and the number of
crossings along R2 side is : (N1sin + N2cos )R2.
Then the hando rate H can be written as
H = (N1cos + N2sin )R1 + (N1sin + N2cos )R2: (3.8)
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Figure 3.5: Hando process with a rectangular cell inclined at an angle .
Now, given the xed area A = R1R2, we need to nd min
H for a given . Replacing
R1 by A
R2
and equating d H
dR1
to zero, we get
R2
1 = A(N1sin + N2cos
N1cos + N2sin
): (3.9)
Similarly, for R2, we get
R2
2 = A(N1cos + N2sin
N1sin + N2cos
): (3.10)
From the above equations, we have H = 2
q
A(N1N2 + (N2
1 + N2
2 )cos sin ) which
means it it minimized at = 0o. Hence min
H = 2
p
AN1N2. Putting the value of
in (3.9) or (3.10), we have R1
R2
= N1
N2
. This has two implications: (i) that hando is
minimized if rectangular cell is aligned with X-Y axis, i.e., = 0o, and, (ii) that the
number of users crossing the cell boundary is inversely proportional to the dimension
of the other side of the cell. The above analysis has been carried out for a simple
square cell and it changes in more complicated way when we consider a hexagonal
cell.
3.5.2 Hando s In Di erent Generations
In 1G analog cellular systems, the signal strength measurements were made by
the BS and in turn supervised by the MSC. The hando s in this generation can
be termed as Network Controlled Hand-O (NCHO). The BS monitors the signal
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strengths of voice channels to determine the relative positions of the subscriber.
The special receivers located on the BS are controlled by the MSC to monitor the
signal strengths of the users in the neighboring cells which appear to be in need
of hando . Based on the information received from the special receivers the MSC
decides whether a hando is required or not. The approximate time needed to make
a hando successful was about 5-10 s. This requires the value of to be in the
order of 6dB to 12dB.
In the 2G systems, the MSC was relieved from the entire operation. In this
generation, which started using the digital technology, hando decisions were mobile
assisted and therefore it is called Mobile Assisted Hand-O (MAHO). In MAHO,
the mobile center measures the power changes received from nearby base stations
and noti es the two BS. Accordingly the two BS communicate and channel transfer
occurs. As compared to 1G, the circuit complexity was increased here whereas the
delay in hando was reduced to 1-5 s. The value of was in the order of 0-5 dB.
However, even this amount of delay could create a communication pause.
In the current 3G systems, the MS measures the power from adjacent BS and
automatically upgrades the channels to its nearer BS. Hence this can be termed as
Mobile Controlled Hand-O (MCHO). When compared to the other generations,
delay during hando is only 100 ms and the value of is around 20 dBm. The
Quality Of Service (QoS) has improved a lot although the complexity of the circuitry
has further increased which is inevitable.
All these types of hando s are usually termed as hard hando as there is a shift
in the channels involved. There is also another kind of hando , called soft hando ,
as discussed below.
Hando in CDMA: In spread spectrum cellular systems, the mobiles share the same
channels in every cell. The MSC evaluates the signal strengths received from di erent
BS for a single user and then shifts the user from one BS to the other without actually
changing the channel. These types of hando s are called as soft hando as there is
no change in the channel.
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3.5.3 Hando Priority
While assigning channels using either FCA or DCA strategy, a guard channel concept
must be followed to facilitate the hando s. This means, a fraction of total available
channels must be kept for hando requests. But this would reduce the carried
tra c and only fewer channels can be assigned for the residual users of a cell. A
good solution to avoid such a dead-lock is to use DCA with hando priority (demand
based allocation).
3.5.4 A Few Practical Problems in Hando Scenario
(a) Di erent speed of mobile users: with the increase of mobile users in urban areas,
microcells are introduced in the cells to increase the capacity (this will be discussed
later in this chapter). The users with high speed frequently crossing the micro-cells
become burdened to MSC as it has to take care of hando s. Several schemes thus
have been designed to handle the simultaneous tra c of high speed and low speed
users while minimizing the hando intervention from the MSC, one of them being
the `Umbrella Cell' approach. This technique provides large area coverage to high
speed users while providing small area coverage to users traveling at low speed. By
using di erent antenna heights and di erent power levels, it is possible to provide
larger and smaller cells at a same location. As illustrated in the Figure 3.6, umbrella
cell is co-located with few other microcells. The BS can measure the speed of the
user by its short term average signal strength over the RVC and decides which cell
to handle that call. If the speed is less, then the corresponding microcell handles
the call so that there is good corner coverage. This approach assures that hando s
are minimized for high speed users and provides additional microcell channels for
pedestrian users.
(b) Cell dragging problem: this is another practical problem in the urban area with
additional microcells. For example, consider there is a LOS path between the MS
and BS1 while the user is in the cell covered by BS2. Since there is a LOS with the
BS1, the signal strength received from BS1 would be greater than that received from
BS2. However, since the user is in cell covered by BS2, hando cannot take place
and as a result, it experiences a lot of interferences. This problem can be solved by
judiciously choosing the hando threshold along with adjusting the coverage area.
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(c) Inter-system hando : if one user is leaving the coverage area of one MSC and is
entering the area of another MSC, then the call might be lost if there is no hando in
this case too. Such a hando is called inter-system hando and in order to facilitate
this, mobiles usually have roaming facility.
3.6 Interference & System Capacity
Susceptibility and interference problems associated with mobile communications
equipment are because of the problem of time congestion within the electromagnetic
spectrum. It is the limiting factor in the performance of cellular systems. This
interference can occur from clash with another mobile in the same cell or because
of a call in the adjacent cell. There can be interference between the base stations
operating at same frequency band or any other non-cellular system's energy leaking
inadvertently into the frequency band of the cellular system. If there is an interference
in the voice channels, cross talk is heard will appear as noise between the users.
The interference in the control channels leads to missed and error calls because of
digital signaling. Interference is more severe in urban areas because of the greater
RF noise and greater density of mobiles and base stations. The interference can be
divided into 2 parts: co-channel interference and adjacent channel interference.
3.6.1 Co-channel interference (CCI)
For the e cient use of available spectrum, it is necessary to reuse frequency bandwidth
over relatively small geographical areas. However, increasing frequency reuse
also increases interference, which decreases system capacity and service quality. The
cells where the same set of frequencies is used are call co-channel cells. Co-channel
interference is the cross talk between two di erent radio transmitters using the same
radio frequency as is the case with the co-channel cells. The reasons of CCI can be
because of either adverse weather conditions or poor frequency planning or overlycrowded
radio spectrum.
If the cell size and the power transmitted at the base stations are same then CCI
will become independent of the transmitted power and will depend on radius of the
cell (R) and the distance between the interfering co-channel cells (D). If D/R ratio
is increased, then the e ective distance between the co-channel cells will increase
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and interference will decrease. The parameter Q is called the frequency reuse ratio
and is related to the cluster size. For hexagonal geometry
Q = D=R =
p
3N: (3.11)
From the above equation, small of `Q' means small value of cluster size `N' and
increase in cellular capacity. But large `Q' leads to decrease in system capacity
but increase in transmission quality. Choosing the options is very careful for the
selection of `N', the proof of which is given in the rst section.
The Signal to Interference Ratio (SIR) for a mobile receiver which monitors the
forward channel can be calculated as
S
I
= S
Pi0
i=1 Ii
(3.12)
where i0 is the number of co-channel interfering cells, S is the desired signal power
from the baseband station and Ii is the interference power caused by the i-th interfering
co-channel base station. In order to solve this equation from power calculations,
we need to look into the signal power characteristics. The average power in the
mobile radio channel decays as a power law of the distance of separation between
transmitter and receiver. The expression for the received power Pr at a distance d
can be approximately calculated as
Pr = P0( d
d0
)􀀀n (3.13)
and in the dB expression as
Pr(dB) = P0(dB) 􀀀 10n log( d
d0
) (3.14)
where P0 is the power received at a close-in reference point in the far eld region at
a small distance do from the transmitting antenna, and `n' is the path loss exponent.
Let us calculate the SIR for this system. If Di is the distance of the i-th interferer
from the mobile, the received power at a given mobile due to i-th interfering cell
is proportional to (Di)􀀀n (the value of 'n' varies between 2 and 4 in urban cellular
systems).
Let us take that the path loss exponent is same throughout the coverage area
and the transmitted power be same, then SIR can be approximated as
S
I
= R􀀀n
Pi0
i=1 D􀀀n
i
(3.15)
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where the mobile is assumed to be located at R distance from the cell center. If
we consider only the rst layer of interfering cells and we assume that the interfering
base stations are equidistant from the reference base station and the distance
between the cell centers is 'D' then the above equation can be converted as
S
I
=
(D=R)n
i0
=
(
p
3N)n
i0
(3.16)
which is an approximate measure of the SIR. Subjective tests performed on AMPS
cellular system which uses FM and 30 kHz channels show that su cient voice quality
can be obtained by SIR being greater than or equal to 18 dB. If we take n=4
, the value of 'N' can be calculated as 6.49. Therefore minimum N is 7. The
above equations are based on hexagonal geometry and the distances from the closest
interfering cells can vary if di erent frequency reuse plans are used.
We can go for a more approximate calculation for co-channel SIR. This is the
example of a 7 cell reuse case. The mobile is at a distance of D-R from 2 closest
interfering cells and approximately D+R/2, D, D-R/2 and D+R distance from other
interfering cells in the rst tier. Taking n = 4 in the above equation, SIR can be
approximately calculated as
S
I
= R􀀀4
2(D 􀀀 R)􀀀4 + (D + R)􀀀4 + (D)􀀀4 + (D + R=2)􀀀4 + (D 􀀀 R=2)􀀀4 (3.17)
which can be rewritten in terms frequency reuse ratio Q as
S
I
=
1
2(Q 􀀀 1)􀀀4 + (Q + 1)􀀀4 + (Q)􀀀4 + (Q + 1=2)􀀀4 + (Q 􀀀 1=2)􀀀4 : (3.18)
Using the value of N equal to 7 (this means Q = 4.6), the above expression yields
that worst case SIR is 53.70 (17.3 dB). This shows that for a 7 cell reuse case the
worst case SIR is slightly less than 18 dB. The worst case is when the mobile is at
the corner of the cell i.e., on a vertex as shown in the Figure 3.6. Therefore N = 12
cluster size should be used. But this reduces the capacity by 7/12 times. Therefore,
co-channel interference controls link performance, which in a way controls frequency
reuse plan and the overall capacity of the cellular system. The e ect of co-channel
interference can be minimized by optimizing the frequency assignments of the base
stations and their transmit powers. Tilting the base-station antenna to limit the
spread of the signals in the system can also be done.
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Figure 3.6: First tier of co-channel interfering cells
3.6.2 Adjacent Channel Interference (ACI)
This is a di erent type of interference which is caused by adjacent channels i.e.
channels in adjacent cells. It is the signal impairment which occurs to one frequency
due to presence of another signal on a nearby frequency. This occurs when imperfect
receiver lters allow nearby frequencies to leak into the passband. This problem is
enhanced if the adjacent channel user is transmitting in a close range compared to
the subscriber's receiver while the receiver attempts to receive a base station on the
channel. This is called near-far e ect. The more adjacent channels are packed into
the channel block, the higher the spectral e ciency, provided that the performance
degradation can be tolerated in the system link budget. This e ect can also occur
if a mobile close to a base station transmits on a channel close to one being used
by a weak mobile. This problem might occur if the base station has problem in
discriminating the mobile user from the "bleed over" caused by the close adjacent
channel mobile.
Adjacent channel interference occurs more frequently in small cell clusters and heavily
used cells. If the frequency separation between the channels is kept large this
interference can be reduced to some extent. Thus assignment of channels is given
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such that they do not form a contiguous band of frequencies within a particular
cell and frequency separation is maximized. E cient assignment strategies are very
much important in making the interference as less as possible. If the frequency factor
is small then distance between the adjacent channels cannot put the interference
level within tolerance limits. If a mobile is 10 times close to the base station than
other mobile and has energy spill out of its passband, then SIR for weak mobile is
approximately
S
I
= 10􀀀n (3.19)
which can be easily found from the earlier SIR expressions. If n = 4, then SIR is
􀀀52 dB. Perfect base station lters are needed when close-in and distant users share
the same cell. Practically, each base station receiver is preceded by a high Q cavity
lter in order to remove adjacent channel interference. Power control is also very
much important for the prolonging of the battery life for the subscriber unit but also
reduces reverse channel SIR in the system. Power control is done such that each
mobile transmits the lowest power required to maintain a good quality link on the
reverse channel.
3.7 Enhancing Capacity And Cell Coverage
3.7.1 The Key Trade-o
Previously, we have seen that the frequency reuse technique in cellular systems
allows for almost boundless expansion of geographical area and the number of mobile
system users who could be accommodated. In designing a cellular layout, the two
parameters which are of great signi cance are the cell radius R and the cluster size
N, and we have also seen that co-channel cell distance D =
p
3NR. In the following,
a brief description of the design trade-o is given, in which the above two parameters
play a crucial role.
The cell radius governs both the geographical area covered by a cell and also
the number of subscribers who can be serviced, given the subscriber density. It is
easy to see that the cell radius must be as large as possible. This is because, every
cell requires an investment in a tower, land on which the tower is placed, and radio
transmission equipment and so a large cell size minimizes the cost per subscriber.
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Eventually, the cell radius is determined by the requirement that adequate signal
to noise ratio be maintained over the coverage area. The SNR is determined by
several factors such as the antenna height, transmitter power, receiver noise gure
etc. Given a cell radius R and a cluster size N, the geographic area covered by a
cluster is
Acluster = NAcell = N3
p
3R2=2: (3.20)
If the total serviced area is Atotal, then the number of clusters M that could be
accommodated is given by
M = Atotal=Acluster = Atotal=(N3
p
3R2=2): (3.21)
Note that all of the available channels N, are reused in every cluster. Hence, to make
the maximum number of channels available to subscribers, the number of clusters
M should be large, which, by Equation (3.21), shows that the cell radius should
be small. However, cell radius is determined by a trade-o : R should be as large
as possible to minimize the cost of the installation per subscriber, but R should
be as small as possible to maximize the number of customers that the system can
accommodate. Now, if the cell radius R is xed, then the number of clusters could be
maximized by minimizing the size of a cluster N. We have seen earlier that the size
of a cluster depends on the frequency reuse ratio Q. Hence, in determining the value
of N, another trade-o is encountered in that N must be small to accommodate
large number of subscribers, but should be su ciently large so as to minimize the
interference e ects.
Now, we focus on the issues regarding system expansion. The history of cellular
phones has been characterized by a rapid growth and expansion in cell subscribers.
Though a cellular system can be expanded by simply adding cells to the geographical
area, the way in which user density can be increased is also important to look at.
This is because it is not always possible to counter the increasing demand for cellular
systems just by increasing the geographical coverage area due to the limitations in
obtaining new land with suitable requirements. We discuss here two methods for
dealing with an increasing subscriber density: Cell Splitting and Sectoring. The
other method, microcell zone concept can treated as enhancing the QoS in a cellular
system.
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The basic idea of adopting the cellular approach is to allow space for the growth
of mobile users. When a new system is deployed, the demand for it is fairly low and
users are assumed to be uniformly distributed over the service area. However, as new
users subscribe to the cellular service, the demand for channels may begin to exceed
the capacity of some base stations. As discussed previously,the number of channels
available to customers (equivalently, the channel density per square kilometer) could
be increased by decreasing the cluster size. However, once a system has been initially
deployed, a system-wide reduction in cluster size may not be necessary since user
density does not grow uniformly in all parts of the geographical area. It might be
that an increase in channel density is required only in speci c parts of the system
to support an increased demand in those areas. Cell-splitting is a technique which
has the capability to add new smaller cells in speci c areas of the system.
3.7.2 Cell-Splitting
Cell Splitting is based on the cell radius reduction and minimizes the need to modify
the existing cell parameters. Cell splitting involves the process of sub-dividing a
congested cell into smaller cells, each with its own base station and a corresponding
reduction in antenna size and transmitting power. This increases the capacity of
a cellular system since it increases the number of times that channels are reused.
Since the new cells have smaller radii than the existing cells, inserting these smaller
cells, known as microcells, between the already existing cells results in an increase
of capacity due to the additional number of channels per unit area. There are few
challenges in increasing the capacity by reducing the cell radius. Clearly, if cells
are small, there would have to be more of them and so additional base stations
will be needed in the system. The challenge in this case is to introduce the new
base stations without the need to move the already existing base station towers.
The other challenge is to meet the generally increasing demand that may vary quite
rapidly between geographical areas of the system. For instance, a city may have
highly populated areas and so the demand must be supported by cells with the
smallest radius. The radius of cells will generally increase as we move from urban to
sub urban areas, because the user density decreases on moving towards sub-urban
areas. The key factor is to add as minimum number of smaller cells as possible
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Figure 3.7: Splitting of congested seven-cell clusters.
wherever an increase in demand occurs. The gradual addition of the smaller cells
implies that, at least for a time, the cellular system operates with cells of more than
one size.
Figure 3.7 shows a cellular layout with seven-cell clusters. Consider that the cells
in the center of the diagram are becoming congested, and cell A in the center has
reached its maximum capacity. Figure also shows how the smaller cells are being
superimposed on the original layout. The new smaller cells have half the cell radius
of the original cells. At half the radius, the new cells will have one-fourth of the area
and will consequently need to support one-fourth the number of subscribers. Notice
that one of the new smaller cells lies in the center of each of the larger cells. If
we assume that base stations are located in the cell centers, this allows the original
base stations to be maintained even in the new system layout. However, new base
stations will have to be added for new cells that do not lie in the center of the larger
cells. The organization of cells into clusters is independent of the cell radius, so that
the cluster size can be the same in the small-cell layout as it was in the large-cell
layout. Also the signal-to-interference ratio is determined by cluster size and not by
cell radius. Consequently, if the cluster size is maintained, the signal-to-interference
ratio will be the same after cell splitting as it was before. If the entire system is
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replaced with new half-radius cells, and the cluster size is maintained, the number
of channels per cell will be exactly as it was before, and the number of subscribers
per cell will have been reduced.
When the cell radius is reduced by a factor, it is also desirable to reduce the
transmitted power. The transmit power of the new cells with radius half that of the
old cells can be found by examining the received power PR at the new and old cell
boundaries and setting them equal. This is necessary to maintain the same frequency
re-use plan in the new cell layout as well. Assume that PT1 and PT2 are the transmit
powers of the larger and smaller base stations respectively. Then, assuming a path
loss index n=4, we have power received at old cell boundary = PT1=R4 and the
power received at new cell boundary = PT2=(R=2)4. On equating the two received
powers, we get PT2 = PT1 / 16. In other words, the transmit power must be reduced
by 12 dB in order to maintain the same S/I with the new system lay-out.
At the beginning of this channel splitting process, there would be fewer channels
in the smaller power groups. As the demand increases, more and more channels need
to be accommodated and hence the splitting process continues until all the larger
cells have been replaced by the smaller cells, at which point splitting is complete
within the region and the entire system is rescaled to have a smaller radius per cell.
If a cellular layout is replaced entirety by a new layout with a smaller cell radius,
the signal-to-interference ratio will not change, provided the cluster size does not
change. Some special care must be taken, however, to avoid co-channel interference
when both large and small cell radii coexist. It turns out that the only way to
avoid interference between the large-cell and small-cell systems is to assign entirely
di erent sets of channels to the two systems. So, when two sizes of cells co-exist in
a system, channels in the old cell must be broken down into two groups, one that
corresponds to larger cell reuse requirements and the other which corresponds to the
smaller cell reuse requirements. The larger cell is usually dedicated to high speed
users as in the umbrella cell approach so as to minimize the number of hand-o s.
Ex. 4: When the AMPS cellular system was rst deployed, the aim of the
system designers was to guarantee coverage. Initially the number of users was not
signi cant. Consequently cells were con gured with an eight-mile radius, and a
12-cell cluster size was chosen. The cell radius was chosen to guarantee a 17 dB
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Figure 3.8: A cell divided into three 120o sectors.
signal-to-noise ratio over 90% of the coverage area. Although a 12-cell cluster size
provided more than adequate co-channel separation to meet a requirement for a
17 dB signal-to-interference ratio in an interference-limited environment, it did not
provide adequate frequency reuse to service an explosively growing customer base.
The system planners reasoned that a subsequent shift to a 7-cell cluster size would
provide an adequate number of channels. It was estimated that a 7-cell cluster
size should provide an adequate 18.7 dB signal-to-interference ratio. The margin,
however, is slim, and the 17 dB signal-to-interference ratio requirement could not
be met over 90 % of the coverage area.
3.7.3 Sectoring
Sectoring is basically a technique which can increase the SIR without necessitating
an increase in the cluster size. Till now, it has been assumed that the base station is
located in the center of a cell and radiates uniformly in all the directions behaving as
an omni-directional antenna. However it has been found that the co-channel interference
in a cellular system may be decreased by replacing a single omni-directional
antenna at the base station by several directional antennas, each radiating within a
speci ed sector. In the Figure 3.8, a cell is shown which has been split into three
120o sectors. The base station feeds three 120o directional antennas, each of which
radiates into one of the three sectors. The channel set serving this cell has also been
divided, so that each sector is assigned one-third of the available number cell of
channels. This technique for reducing co-channel interference wherein by using suit-
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Figure 3.9: A seven-cell cluster with 60o sectors.
able directional antennas, a given cell would receive interference and transmit with
a fraction of available co-channel cells is called 'sectoring'. In a seven-cell-cluster
layout with 120o sectored cells, it can be easily understood that the mobile units in
a particular sector of the center cell will receive co-channel interference from only
two of the rst-tier co-channel base stations, rather than from all six. Likewise, the
base station in the center cell will receive co-channel interference from mobile units
in only two of the co-channel cells. Hence the signal to interference ratio is now
modi ed to
S
I
=
(
p
3N)n
2
(3.22)
where the denominator has been reduced from 6 to 2 to account for the reduced
number of interfering sources. Now, the signal to interference ratio for a seven-cell
cluster layout using 120o sectored antennas can be found from equation (3.24) to be
23.4 dB which is a signi cant improvement over the Omni-directional case where the
worst-case S/I is found to be 17 dB (assuming a path-loss exponent, n=4). Some
cellular systems divide the cells into 60o sectors. Similar analysis can be performed
on them as well.
Ex. 5: A cellular system having a seven-cell cluster layout with omni-directional
antennas has been performing satisfactorily for a required signal to interference ratio
of 15 dB. However due to the need for increasing the number of available channels, a
60o sectoring of the cells has been introduced. By what percentage can the number
of channels Ntotal be increased assuming a path-loss component n=4?
Solution: The seven-cell cluster layout with 60o sectoring is shown in the Figure 3.9.
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It is easy to see that the shaded region in the center receives interference from just
one rst-tier cell and hence the signal to interference ratio can be obtained suitably
as
S
I
=
(
p
3N)n
1
=
(
p
(3)(7))4
1
= 26:4dB: (3.23)
Since the SIR exceeds 15 dB, one can try reducing the cluster size from seven to
four. Now, the SIR for this reduced cluster size layout can be found to be
S
I
=
(
p
3N)n
1
=
(
p
(3)(4))4
1
= 21:6dB: (3.24)
The S/I ratio is still above the requirement and so a further reduction in the cell
cluster size is possible. For a 3-cell cluster layout, there are two interfering sources
and hence the S/I ratio is found to be
S
I
=
(
p
3N)n
1
=
(
p
33)4
2
= 16:07dB: (3.25)
This is just above the adequate S/I ratio and further reduction in cluster size is
not possible. So, a 3-cluster cell layout could be used for meeting the growth requirements.
Thus, when the cluster size is reduced from 7 to 3, the total number of
channels increased by a factor of 7/3.
The calculations in the above example are actually an idealization for several
reasons. Firstly, practical antennas have side lobes and cannot be used to focus a
transmitted beam into a perfect 120o sector or 60o sector. Due to this, additional
interference will be introduced. Next, it is also a cause of concern that a given
number of channels are not able to support as many subscribers when the pool
of channels is divided into small groups. This is due to a reduction in Trunking
E ciency, a term which will be explained later on. Because sectoring involves using
more than one antenna per base station, the available channels in the cell are divided
and dedicated to a speci c antenna. This breaks the available set of channels into
smaller sets, thus reducing the trunking e ciency. Moreover, dividing a cell into
sectors requires that a call in progress will have to be handed o (that is, assigned
a new channel) when a mobile unit travels into a new sector. This increases the
complexity of the system and also the load on the mobile switching center/base
station.
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3.7.4 Microcell Zone Concept
The increased number of hando s required when sectoring is employed results in an
increased load on the switching and control link elements of the mobile system. To
overcome this problem, a new microcell zone concept has been proposed. As shown
in Figure 3.10, this scheme has a cell divided into three microcell zones, with each
of the three zone sites connected to the base station and sharing the same radio
equipment. It is necessary to note that all the microcell zones, within a cell, use the
same frequency used by that cell; that is no handovers occur between microcells.
Thus when a mobile user moves between two microcell zones of the cell, the BS
simply switches the channel to a di erent zone site and no physical re-allotment of
channel takes place.
Locating the mobile unit within the cell: An active mobile unit sends a signal to all
zone sites, which in turn send a signal to the BS. A zone selector at the BS uses that
signal to select a suitable zone to serve the mobile unit - choosing the zone with the
strongest signal.
Base Station Signals: When a call is made to a cellular phone, the system already
knows the cell location of that phone. The base station of that cell knows in which
zone, within that cell, the cellular phone is located. Therefore when it receives the
signal, the base station transmits it to the suitable zone site. The zone site receives
the cellular signal from the base station and transmits that signal to the mobile
phone after ampli cation. By con ning the power transmitted to the mobile phone,
co-channel interference is reduced between the zones and the capacity of system is
increased.
Bene ts of the micro-cell zone concept: 1) Interference is reduced in this case as
compared to the scheme in which the cell size is reduced.
2) Hando s are reduced (also compared to decreasing the cell size) since the microcells
within the cell operate at the same frequency; no handover occurs when the
mobile unit moves between the microcells.
3) Size of the zone apparatus is small. The zone site equipment being small can be
mounted on the side of a building or on poles.
4) System capacity is increased. The new microcell knows where to locate the mobile
unit in a particular zone of the cell and deliver the power to that zone. Since
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Figure 3.10: The micro-cell zone concept.
the signal power is reduced, the microcells can be closer and result in an increased
system capacity. However, in a microcellular system, the transmitted power to a
mobile phone within a microcell has to be precise; too much power results in interference
between microcells, while with too little power the signal might not reach
the mobile phone.This is a drawback of microcellular systems, since a change in the
surrounding (a new building, say, within a microcell) will require a change of the
transmission power.
3.8 Trunked Radio System
In the previous sections, we have discussed the frequency reuse plan, the design
trade-o s and also explored certain capacity expansion techniques like cell-splitting
and sectoring. Now, we look at the relation between the number of radio channels
a cell contains and the number of users a cell can support. Cellular systems use
the concept of trunking to accommodate a large number of users in a limited radio
spectrum. It was found that a central o ce associated with say, 10,000 telephones
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requires about 50 million connections to connect every possible pair of users. However,
a worst case maximum of 5000 connections need to be made among these
telephones at any given instant of time, as against the possible 50 million connections.
In fact, only a few hundreds of lines are needed owing to the relatively short
duration of a call. This indicates that the resources are shared so that the number of
lines is much smaller than the number of possible connections. A line that connects
switching o ces and that is shared among users on an as-needed basis is called a
trunk.
The fact that the number of trunks needed to make connections between o ces
is much smaller than the maximum number that could be used suggests that at
times there might not be su cient facilities to allow a call to be completed. A call
that cannot be completed owing to a lack of resources is said to be blocked. So one
important to be answered in mobile cellular systems is: How many channels per cell
are needed in a cellular telephone system to ensure a reasonably low probability that
a call will be blocked?
In a trunked radio system, a channel is allotted on per call basis. The performance
of a radio system can be estimated in a way by looking at how e ciently the
calls are getting connected and also how they are being maintained at hando s.
Some of the important factors to take into consideration are (i) Arrival statistics,
(ii)Service statistics, (iii)Number of servers/channels.
Let us now consider the following assumptions for a bu erless system handling 'L'
users as shown in Figure 3.11:
(i) The number of users L is large when compared to 1.
(ii) Arrival statistics is Poisson distributed with a mean parameter .
(iii) Duration of a call is exponentially distributed with a mean rate 1.
(iv) Residence time of each user is exponentially distributed with a rate parameter
2.
(v) The channel holding rate therefore is exponentially distributed with a parameter
= 1 + 2.
(vi) There is a total of 'J' number of channels (J L).
To analyze such a system, let us recapitulate a queuing system in brief. Consider an
M/M/m/m system which is an m-server loss system. The name M/M/m/m re
ects
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Figure 3.11: The bu erless J-channel trunked radio system.
Figure 3.12: Discrete-time Markov chain for the M/M/J/J trunked radio system.
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standard queuing theory nomenclature whereby:
(i) the rst letter indicates the nature of arrival process(e.g. M stands for memoryless
which here means a Poisson process).
(ii) the second letter indicates the nature of probability distribution of service
times.(e.g M stands for exponential distribution). In all cases,successive inter arrival
times and service times are assumed to be statistically independent of each
other.
(iii) the third letter indicates the number of servers.
(iv) the last letter indicates that if an arrival nds all 'm' users to be busy, then it
will not enter the system and is lost.
In view of the above, the bu erless system as shown in Figure 3.11 can be modeled
as M/M/J/J system and the discrete-time Markov chain of this system is shown in
Figure 3.12.
Trunking mainly exploits the statistical behavior of users so that a xed number
of channels can be used to accommodate a large, random user community. As the
number of telephone lines decrease, it becomes more likely that all channels are
busy for a particular user. As a result, the call gets rejected and in some systems,
a queue may be used to hold the caller's request until a channel becomes available.
In the telephone system context the term Grade of Service (GoS) is used to mean
the probability that a user's request for service will be blocked because a required
facility, such as a trunk or a cellular channel, is not available. For example, a GoS of
2 % implies that on the average a user might not be successful in placing a call on 2
out of every 100 attempts. In practice the blocking frequency varies with time. One
would expect far more call attempts during business hours than during the middle of
the night. Telephone operating companies maintain usage records and can identify a
"busy hour", that is, the hour of the day during which there is the greatest demand
for service. Typically, telephone systems are engineered to provide a speci ed grade
of service during a speci ed busy hour.
User calling can be modeled statistically by two parameters: the average number
of call requests per unit time user and the average holding time H. The parameter
user is also called the average arrival rate, referring to the rate at which calls from
a single user arrive. The average holding time is the average duration of a call. The
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product:
Auser = userH (3.26)
that is, the product of the average arrival rate and the average holding time{is called
the o ered tra c intensity or o ered load. This quantity represents the average
tra c that a user provides to the system. O ered tra c intensity is a quantity that
is traditionally measured in Erlangs. One Erlang represents the amount of tra c
intensity carried by a channel that is completely occupied. For example, a channel
that is occupied for thirty minutes during an hour carries 0.5 Erlang of tra c.
Call arrivals or requests for service are modeled as a Poisson random process. It
is based on the assumption that there is a large pool of users who do not cooperate
in deciding when to place calls. Holding times are very well predicted using an
exponential probability distribution. This implies that calls of long duration are
much less frequent than short calls. If the tra c intensity o ered by a single user is
Auser, then the tra c intensity o ered by N users is A = NAuser. The purpose of
the statistical model is to relate the o ered tra c intensity A, the grade of service
Pb, and the number of channels or trunks C needed to maintain the desired grade
of service.
Two models are widely used in tra c engineering to represent what happens
when a call is blocked. The blocked calls cleared model assumes that when a channel
or trunk is not available to service an arriving call, the call is cleared from the
system. The second model is known as blocked calls delayed. In this model a call
that cannot be serviced is placed on a queue and will be serviced when a channel or
trunk becomes available.
Use of the blocked-calls-cleared statistical model leads to the Erlang B formula
that relates o ered tra c intensity A, grade of service Pb, and number of channels
K. The Erlang B formula is:
Pb = AK=K!
PK
n=0 An=n!
(3.27)
When the blocked-calls-delayed model is used, the "grade of service" refers to the
probability that a call will be delayed. In this case the statistical model leads to the
Erlang C formula,
P[delay] = AK=[(K 􀀀 A)(K 􀀀 1)]!
AK=[(K 􀀀 A)(K 􀀀 1)]! +
PK
n=0 An=n!
: (3.28)
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Ex. 6: In a certain cellular system, an average subscriber places two calls per
hour during a busy hour and the average holding time is 3 min. Each cell has 100
channels. If the blocked calls are cleared, how many subscribers can be serviced by
each cell at 2 % GoS?
Solution: Using Erlang B table, it can be seen that for C = 100 and GoS = Pb = 2%,
the total o ered load A=87.972 Erlangs. Since an individual subscriber o ers a load
of Auser = (2 calls / 60 min)3 min = 0.1 Erlang, the maximum number of subscribers
served is
N = A=Auser = 87:972=0:1 880: (3.29)
Ex. 4: In the previous example, suppose that the channels have been divided into
two groups of 50 channels each. Each subscriber is assigned to a group and can be
served only by that group. How many subscribers can be served by the two group
cell?
Solution: Using the Erlang B table with C = 50 and GOS = Pb = 2%, the total
o ered load per group is
A = 40:255Erlangs (3.30)
Thus the maximum number of users per group is
Ngroup = A=Auser 403: (3.31)
Thus, counting both the groups, maximum number of users in the two group cell is
806.
The above example indicates that the number of subscribers that can be supported
by a given number of channels decreases as the pool of channels is sub-divided.
We can express this in terms of the trunking e ciency, de ned as the carrier load
per channel, that is,
= (1 􀀀 Pb)A=C: (3.32)
This explains why the sectoring of a cell into either 120o or 60o sectors reduces
the trunking e ciency of the system. Thus the system growth due to sectoring is
impacted by trunking e ciency considerations.
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3.9 References
1. T. S. Rappaport, Wireless Communications: Principles and Practice, 2nd ed.
Singapore: Pearson Education, Inc., 2002.
2. K. Feher, Wireless Digital Communications: Modulation and Spread Spectrum
Applications. Upper Saddle River, NJ: Prentice Hall, 1995.
3. S. Haykin and M. Moher, Modern Wireless Communications. Singapore: Pearson
Education, Inc., 2002.
4. J. W. Mark and W. Zhuang, Wireless Communications and Networking. New
Delhi: PHI, 2005.
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Chapter 4
Free Space Radio Wave
Propagation
4.1 Introduction
There are two basic ways of transmitting an electro-magnetic (EM) signal, through a
guided medium or through an unguided medium. Guided mediums such as coaxial
cables and ber optic cables, are far less hostile toward the information carrying
EM signal than the wireless or the unguided medium. It presents challenges and
conditions which are unique for this kind of transmissions. A signal, as it travels
through the wireless channel, undergoes many kinds of propagation e ects such as
re
ection, di raction and scattering, due to the presence of buildings, mountains and
other such obstructions. Re
ection occurs when the EM waves impinge on objects
which are much greater than the wavelength of the traveling wave. Di raction
is a phenomena occurring when the wave interacts with a surface having sharp
irregularities. Scattering occurs when the medium through the wave is traveling
contains objects which are much smaller than the wavelength of the EM wave.
These varied phenomena's lead to large scale and small scale propagation losses. Due
to the inherent randomness associated with such channels they are best described
with the help of statistical models. Models which predict the mean signal strength
for arbitrary transmitter receiver distances are termed as large scale propagation
models. These are termed so because they predict the average signal strength for
large Tx-Rx separations, typically for hundreds of kilometers.
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Figure 4.1: Free space propagation model, showing the near and far elds.
4.2 Free Space Propagation Model
Although EM signals when traveling through wireless channels experience fading
e ects due to various e ects, but in some cases the transmission is with a direct
line of sight such as in satellite communication. Free space model predicts that
the received power decays as negative square root of the distance. Friis free space
equation is given by
Pr(d) = PtGtGr 2
(4 )2d2L
(4.1)
where Pt is the transmitted power, Pr(d) is the received power, Gt is the transmitter
antenna gain, Gr is the receiver antenna gain, d is the Tx-Rx separation and L is the
system loss factor depended upon line attenuation, lter losses and antenna losses
and not related to propagation. The gain of the antenna is related to the e ective
aperture of the antenna which in turn is dependent upon the physical size of the
antenna as given below
G = 4 Ae= 2: (4.2)
The path loss, representing the attenuation su ered by the signal as it travels
through the wireless channel is given by the di erence of the transmitted and received
power in dB and is expressed as:
PL(dB) = 10 log Pt=Pr: (4.3)
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The elds of an antenna can broadly be classi ed in two regions, the far eld and
the near eld. It is in the far eld that the propagating waves act as plane waves
and the power decays inversely with distance. The far eld region is also termed
as Fraunhofer region and the Friis equation holds in this region. Hence, the Friis
equation is used only beyond the far eld distance, df , which is dependent upon the
largest dimension of the antenna as
df = 2D2= : (4.4)
Also we can see that the Friis equation is not de ned for d=0. For this reason, we
use a close in distance, do, as a reference point. The power received, Pr(d), is then
given by:
Pr(d) = Pr(do)(do=d)2: (4.5)
Ex. 1: Find the far eld distance for a circular antenna with maximum dimension
of 1 m and operating frequency of 900 MHz.
Solution: Since the operating frequency f = 900 Mhz, the wavelength
=
3 108m=s
900 106Hz
m
. Thus, with the largest dimension of the antenna, D=1m, the far eld distance is
df =
2D2

=
2(1)2
0:33
= 6m
.
Ex. 2: A unit gain antenna with a maximum dimension of 1 m produces 50 W
power at 900 MHz. Find (i) the transmit power in dBm and dB, (ii) the received
power at a free space distance of 5 m and 100 m.
Solution:
(i) Tx power = 10log(50) = 17 dB = (17+30) dBm = 47 dBm
(ii) df = 2 D2
= 2 12
1=3 = 6m
Thus the received power at 5 m can not be calculated using free space distance
formula.
At 100 m ,
PR = PTGTGR 2
4 d2
=
50 1 (1=3)2
4 1002
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= 3:5 10􀀀3mW
PR(dBm) = 10logPr(mW) = 􀀀24:5dBm
4.3 Basic Methods of Propagation
Re
ection, di raction and scattering are the three fundamental phenomena that
cause signal propagation in a mobile communication system, apart from LoS communication.
The most important parameter, predicted by propagation models based
on above three phenomena, is the received power. The physics of the above phenomena
may also be used to describe small scale fading and multipath propagation.
The following subsections give an outline of these phenomena.
4.3.1 Re
ection
Re
ection occurs when an electromagnetic wave falls on an object, which has very
large dimensions as compared to the wavelength of the propagating wave. For example,
such objects can be the earth, buildings and walls. When a radio wave falls
on another medium having di erent electrical properties, a part of it is transmitted
into it, while some energy is re
ected back. Let us see some special cases. If the
medium on which the e.m. wave is incident is a dielectric, some energy is re
ected
back and some energy is transmitted. If the medium is a perfect conductor, all
energy is re
ected back to the rst medium. The amount of energy that is re
ected
back depends on the polarization of the e.m. wave.
Another particular case of interest arises in parallel polarization, when no re-

ection occurs in the medium of origin. This would occur, when the incident angle
would be such that the re
ection coe cient is equal to zero. This angle is the
Brewster's angle. By applying laws of electro-magnetics, it is found to be
sin( B) =
r
1
1 + 2
: (4.6)
Further, considering perfect conductors, the electric eld inside the conductor is
always zero. Hence all energy is re
ected back. Boundary conditions require that
i = r (4.7)
and
Ei = Er (4.8)
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for vertical polarization, and
Ei = 􀀀Er (4.9)
for horizontal polarization.
4.3.2 Di raction
Di raction is the phenomenon due to which an EM wave can propagate beyond the
horizon, around the curved earth's surface and obstructions like tall buildings. As
the user moves deeper into the shadowed region, the received eld strength decreases.
But the di raction eld still exists an it has enough strength to yield a good signal.
This phenomenon can be explained by the Huygen's principle, according to
which, every point on a wavefront acts as point sources for the production of secondary
wavelets, and they combine to produce a new wavefront in the direction of
propagation. The propagation of secondary wavelets in the shadowed region results
in di raction. The eld in the shadowed region is the vector sum of the electric eld
components of all the secondary wavelets that are received by the receiver.
4.3.3 Scattering
The actual received power at the receiver is somewhat stronger than claimed by the
models of re
ection and di raction. The cause is that the trees, buildings and lampposts
scatter energy in all directions. This provides extra energy at the receiver.
Roughness is tested by a Rayleigh criterion, which de nes a critical height hc of
surface protuberances for a given angle of incidence i, given by,
hc =
8sin i
: (4.10)
A surface is smooth if its minimum to maximum protuberance h is less than hc,
and rough if protuberance is greater than hc. In case of rough surfaces, the surface
re
ection coe cient needs to be multiplied by a scattering loss factor S, given by
S = exp(􀀀8( hsin i

)2) (4.11)
where h is the standard deviation of the Gaussian random variable h. The following
result is a better approximation to the observed value
S = exp(􀀀8( hsin i

)2)I0[􀀀8( hsin i

)2] (4.12)
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Figure 4.2: Two-ray re
ection model.
which agrees very well for large walls made of limestone. The equivalent re
ection
coe cient is given by,
􀀀rough = S􀀀: (4.13)
4.4 Two Ray Re
ection Model
Interaction of EM waves with materials having di erent electrical properties than
the material through which the wave is traveling leads to transmitting of energy
through the medium and re
ection of energy back in the medium of propagation.
The amount of energy re
ected to the amount of energy incidented is represented
by Fresnel re
ection coe cient 􀀀, which depends upon the wave polarization, angle
of incidence and frequency of the wave. For example, as the EM waves can not pass
through conductors, all the energy is re
ected back with angle of incidence equal to
the angle of re
ection and re
ection coe cient 􀀀 = 􀀀1. In general, for parallel and
perpendicular polarizations, 􀀀 is given by:
􀀀jj = Er=Ei = 2 sin t 􀀀 1 sin i= 2 sin t + 1 sin i (4.14)
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􀀀? = Er=Ei = 2 sin i 􀀀 1 sin t= 2 sin i + 1 sin t: (4.15)
Seldom in communication systems we encounter channels with only LOS paths and
hence the Friis formula is not a very accurate description of the communication link.
A two-ray model, which consists of two overlapping waves at the receiver, one direct
path and one re
ected wave from the ground gives a more accurate description as
shown in Figure 4.2. A simple addition of a single re
ected wave shows that power
varies inversely with the forth power of the distance between the Tx and the Rx.
This is deduced via the following treatment. From Figure 4.2, the total transmitted
and received electric elds are
ETOT
T = Ei + ELOS; (4.16)
ETOT
R = Eg + ELOS: (4.17)
Let E0 is the free space electric eld (in V/m) at a reference distance d0. Then
E(d; t) = E0d0
d
cos(!ct 􀀀 ) (4.18)
where
= !c
d
c
(4.19)
and d > d0. The envelop of the electric eld at d meters from the transmitter at
any time t is therefore
jE(d; t)j = E0d0
d
: (4.20)
This means the envelop is constant with respect to time.
Two propagating waves arrive at the receiver, one LOS wave which travels a
distance of d
0 and another ground re
ected wave, that travels d
00 . Mathematically,
it can be expressed as:
E(d
0
; t) = E0d0
d0 cos(!ct 􀀀
0) (4.21)
where

0 = !c
d
0
c
(4.22)
and
E(d
00
; t) = E0d0
d00 cos(!ct 􀀀
00) (4.23)
where

00 = !c
d
00
c
: (4.24)
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Figure 4.3: Phasor diagram of electric
elds.
Figure 4.4: Equivalent phasor diagram of
Figure 4.3.
According to the law of re
ection in a dielectric, i = 0 and Eg = 􀀀Ei which means
the total electric eld,
Et = Ei + Eg = Ei(1 + 􀀀): (4.25)
For small values of i, re
ected wave is equal in magnitude and 180o out of phase
with respect to incident wave. Assuming perfect horizontal electric eld polarization,
i.e.,
􀀀? = 􀀀1 =) Et = (1 􀀀 1)Ei = 0; (4.26)
the resultant electric eld is the vector sum of ELOS and Eg. This implies that,
ETOT
R = jELOS + Egj: (4.27)
It can be therefore written that
ETOT
R (d; t) = E0d0
d0 cos(!ct 􀀀
0) + (􀀀1)E0d0
d00 cos(!ct 􀀀
00) (4.28)
In such cases, the path di erence is
= d
00
􀀀 d
0 =
q
(ht + hr)2 + d2 􀀀
q
(ht 􀀀 hr)2 + d2: (4.29)
However, when T-R separation distance is very large compared to (ht + hr), then

2hthr
d
(4.30)
Ex 3: Prove the above two equations, i.e., equation (4.29) and (4.30).
Once the path di erence is known, the phase di erence is
=
2

=
!c

(4.31)
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and the time di erence,
d =

c
=
2 fc
: (4.32)
When d is very large, then becomes very small and therefore ELOS and Eg are
virtually identical with only phase di erence,i.e.,
j
E0d0
d
j j
E0d0
d0 j j
E0d0
d00 j: (4.33)
Say, we want to evaluate the received E- eld at any t = d
00
c . Then,
ETOT
R (d; t = d
00
c
) = E0d0
d0 cos(!c
d
00
c
􀀀 !c
d
0
c
) 􀀀
E0d0
d00 cos(!c
d
00
c
􀀀 !c
d
00
c
) (4.34)
= E0d0
d0 cos(
!c
c
) 􀀀
E0d0
d00 cos(0o) (4.35)
= E0d0
d0
6 􀀀
E0d0
d00 (4.36)

E0d0
d
(6 􀀀 1): (4.37)
Using phasor diagram concept for vector addition as shown in Figures 4.3 and 4.4,
we get
jETOT
R (d)j =
s
(E0d0
d
+ E0d0
d
cos( ))2 + (E0d0
d
sin( ))2 (4.38)
= E0d0
d
q
(cos( ) 􀀀 1)2 + sin2( ) (4.39)
= E0d0
d
p
2 􀀀 2cos (4.40)
= 2E0d0
d
sin(
2
): (4.41)
For
2 < 0:5rad, sin(
2 )
2 . Using equation (4.31) and further equation (4.30),
we can then approximate that
sin(
2
)


=
2 hthr
d
< 0:5rad: (4.42)
This raises the wonderful concept of `cross-over distance' dc, de ned as
d > dc =
20 hthr
5
=
4 hthr

: (4.43)
The corresponding approximate received electric eld is
ETOT
R (d) 2E0d0
d
2 hthr
d
= k
hthr
d2 : (4.44)
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Therefore, using equation (4.43) in (4.1), we get the received power as
Pr = PtGtGrh2
t h2
r
Ld4 : (4.45)
The cross-over distance shows an approximation of the distance after which the
received power decays with its fourth order. The basic di erence between equation
(4.1) and (4.45) is that when d < dc, equation (4.1) is su cient to calculate the
path loss since the two-ray model does not give a good result for a short distance
due to the oscillation caused by the constructive and destructive combination of the
two rays, but whenever we distance crosses the `cross-over distance', the power falls
o rapidly as well as two-ray model approximation gives better result than Friis
equation.
Observations on Equation (4.45): The important observations from this
equation are:
1. This equation gives fair results when the T-R separation distance crosses the
cross-over distance.
1. In that case, the power decays as the fourth power of distance
Pr(d) = K
d4 ; (4.46)
with K being a constant.
2. Path loss is independent of frequency (wavelength).
3. Received power is also proportional to h2
t and h2
r , meaning, if height of any of the
antennas is increased, received power increases.
4.5 Di raction
Di raction is the phenomena that explains the digression of a wave from a straight
line path, under the in
uence of an obstacle, so as to propagate behind the obstacle.
It is an inherent feature of a wave be it longitudinal or transverse. For e.g the
sound can be heard in a room, where the source of the sound is another room
without having any line of sight. The similar phenomena occurs for light also but
the di racted light intensity is not noticeable. This is because the obstacle or slit
need to be of the order of the wavelength of the wave to have a signi cant e ect.
Thus radiation from a point source radiating in all directions can be received at any
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Figure 4.5: Huygen's secondary wavelets.
point, even behind an obstacle (unless it is not completely enveloped by it), as shown
in Figure 4.5. Though the intensity received gets smaller as receiver is moved into the
shadowed region. Di raction is explained by Huygens-Fresnel principle which states
that all points on a wavefront can be considered as the point source for secondary
wavelets which form the secondary wavefront in the direction of the prorogation.
Normally, in absence of an obstacle, the sum of all wave sources is zero at a point
not in the direct path of the wave and thus the wave travels in the straight line. But
in the case of an obstacle, the e ect of wave source behind the obstacle cannot be
felt and the sources around the obstacle contribute to the secondary wavelets in the
shadowed region, leading to bending of wave. In mobile communication, this has a
great advantage since, by di raction (and scattering, re
ection), the receiver is able
to receive the signal even when not in line of sight of the transmitter. This we show
in the subsection given below.
4.5.1 Knife-Edge Di raction Geometry
As shown in Figure 4.6, consider that there's an impenetrable obstruction of hight
h at a distance of d1 from the transmitter and d2 from the receiver. The path
di erence between direct path and the di racted path is
=
q
d2
1 + h2 +
q
d2
2 + h2 􀀀 (d1 + d2) (4.47)
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Figure 4.6: Di raction through a sharp edge.
which can be further simpli ed as
= d1(1 + h2=2d2
1 ) + d2(1 + h2=2d2
2) 􀀀 (d1 + d2)
= h2=(2d1) + h2=(2d2) = h2(d1 + d2)=(2d1d2): (4.48)
Thus the phase di erence equals
= 2 = = 2 h2(d1 + d2)= 2(d1d2): (4.49)
With the following considerations that
= +
 (4.50)
and
tan (4.51)
we can write,
tan = tan + tan
 = h=d1 + h=d2 = h(d1 + d2)=d1d2: (4.52)
In order to normalize this, we usually use a Fresnel-Kircho di raction parameter
v, expressed as
v = h
q
2(d1 + d2)=( d1d2) =
q
(2d1d2)=( (d1 + d2)) (4.53)
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Figure 4.7: Fresnel zones.
and therefore the phase di erence becomes
= v2=2: (4.54)
From this, we can observe that: (i) phase di erence is a function of the height of
the obstruction, and also, (ii) phase di erence is a function of the position of the
obstruction from transmitter and receiver.
4.5.2 Fresnel Zones: the Concept of Di raction Loss
As mentioned before, the more is the object in the shadowed region greater is the
di raction loss of the signal. The e ect of di raction loss is explained by Fresnel
zones as a function of the path di erence. The successive Fresnel zones are limited
by the circular periphery through which the path di erence of the secondary waves
is n =2 greater than total length of the LOS path, as shown in Figure 4.7. Thus
successive Fresnel zones have phase di erence of which means they alternatively
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provide constructive and destructive interference to the received the signal. The
radius of the each Fresnel zone is maximum at middle of transmitter and receiver
(i.e. when d1 = d2 ) and decreases as moved to either side. It is seen that the loci
of a Fresnel zone varied over d1 and d2 forms an ellipsoid with the transmitter and
receiver at its focii. Now, if there's no obstruction, then all Fresnel zones result in
only the direct LOS prorogation and no di raction e ects are observed. But if an
obstruction is present, depending on its geometry, it obstructs contribution from
some of the secondary wavelets, resulting in di raction and also the loss of energy,
which is the vector sum of energy from unobstructed sources. please note that height
of the obstruction can be positive zero and negative also. The di raction losses are
minimum as long as obstruction doesn't block volume of the 1st Fresnel zone. As a
rule of thumb, di raction e ects are negligible beyond 55% of 1st Fresnel zone.
Ex 4: Calculate the rst Fresnel zone obstruction height maximum for f = 800
MHz.
Solution:
= c
f
=
3 108
8 102 106 =
3
8m
H =
q
(d1+d2)
d1+d2
H1 =
r
3
8 250 250
500 = 6:89m
Thus H1 = 10 + 6:89 = 16:89m
(b)
H2 =
s
3
8 100 400
500
= 10
q
(0:3) = 5:48m
Thus
H2 = 10 + 5:6 = 15:48m
. To have good power strength, obstacle should be within the 60% of the rst fresnel
zone.
Ex 5: Given f=900 MHz, d1 = d2 = 1 km, h = 25m, where symbols have usual
meaning. Compute the di raction loss. Also nd out in which Fresnel zone the tip
of the obstruction lies.
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Figure 4.8: Knife-edge Di raction Model
Given,
Gd(dB) = 20 log(0:5 􀀀 0:62v) 􀀀 1 < v <= 0
Gd(dB) = 20 log(0:225=v) v > 2:24
Solution:
v = h
s
2(d1 + d2)
d1d2
= 25
s
2 2000
1
310
= 2:74
Gd(dB) = 20 log( 225
v ) = 􀀀21:7dB
Since loss = -Gd (dB) = 21.7 dB
n =
(2:74)2
2
= 3:5
Thus n=4.
4.5.3 Knife-edge di raction model
Knife-edge di raction model is one of the simplest di raction model to estimate the
di raction loss. It considers the object like hill or mountain as a knife edge sharp
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object. The electric eld strength, Ed of a knife-edge di racted wave is given by
Ed=Eo = F(v) = (1 + j)=2
Z 1
v
(exp((􀀀j t2)=2)dt: (4.55)
The di raction gain due to presence of knife edge can be given as
Gd(db) = 20logjF(v)j (4.56)
Gd(db) = 0v <= 􀀀1 (4.57)
Gd(db) = 20log(0:5 􀀀 0:62) 􀀀 1 <= v <= 0 (4.58)
Gd(db) = 20log(0:5exp(􀀀0:95v)) 0 <= v <= 1 (4.59)
Gd(db) = 20log(0:4 􀀀 sqrt(0:1184 􀀀 (0:38 􀀀 0:1v2))) 1 <= v <= 2:4 (4.60)
Gd(db) = 20log(0:225=v) v > 2:4 (4.61)
When there are more than one obstruction, then the equivalent model can be found
by one knife-edge di raction model as shown in Figure 4.8.
4.6 Link Budget Analysis
4.6.1 Log-distance Path Loss Model
According to this model the received power at distance d is given by,
PL(d)( d
d0
)n =) PL(dB) = PL(d0) + 10nlog( d
d0
) (4.62)
The value of n varies with propagation environments. The value of n is 2 for free
space. The value of n varies from 4 to 6 for obstruction of building, and 3 to 5 for
urban scenarios. The important factor is to select the correct reference distance d0.
For large cell area it is 1 Km, while for micro-cell system it varies from 10m-1m.
Limitations:
Surrounding environmental clutter may be di erent for two locations having
the same transmitter to receiver separation. Moreover it does not account for the
shadowing e ects.
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4.6.2 Log Normal Shadowing
The equation for the log normal shadowing is given by,
PL(dB) = PL(dB) + X = PL(d0) + 10nlog( d
d0
) + X (4.63)
where X is a zero mean Gaussian distributed random variable in dB with standard
deviation also in dB. In practice n and values are computed from measured
data.
Average received power
The `Q' function is given by,
Q(z) = 0:5(1 􀀀 erf( z
p
2
)) (4.64)
and
Q(z) = 1 􀀀 Q(􀀀z) (4.65)
So the probability that the received signal level (in dB) will exceed a certain value

 is
P(Pd >
) = Q(
 􀀀 Pr

): (4.66)
4.7 Outdoor Propagation Models
There are many empirical outdoor propagation models such as Longley-Rice model,
Durkin's model, Okumura model, Hata model etc. Longley-Rice model is the most
commonly used model within a frequency band of 40 MHz to 100 GHz over di erent
terrains. Certain modi cations over the rudimentary model like an extra urban
factor (UF) due to urban clutter near the reciever is also included in this model.
Below, we discuss some of the outdoor models, followed by a few indoor models too.
4.7.1 Okumura Model
The Okumura model is used for Urban Areas is a Radio propagation model that is
used for signal prediction.The frequency coverage of this model is in the range of
200 MHz to 1900 MHz and distances of 1 Km to 100 Km.It can be applicable for
base station e ective antenna heights (ht) ranging from 30 m to 1000 m.
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Okumura used extensive measurements of base station-to-mobile signal attenuation
throughout Tokyo to develop a set of curves giving median attenuation relative
to free space (Amu) of signal propogation in irregular terrain. The empirical pathloss
formula of Okumura at distance d parameterized by the carrier frequency fc is
given by
PL(d)dB = L(fc; d) + Amu(fc; d) 􀀀 G(ht) 􀀀 G(hr) 􀀀 GAREA (4.67)
where L(fc; d) is free space path loss at distance d and carrier frequency fc, Amu(fc; d)
is the median attenuation in addition to free-space path loss across all environments,G(ht)
is the base station antenna height gain factor,G(hr) is the mobile antenna height gain
factor,GAREA is the gain due to type of environment. The values of Amu(fc; d) and
GAREA are obtained from Okumura's empirical plots. Okumura derived empirical
formulas for G(ht) and G(hr) as follows:
G(ht) = 20 log10(ht=200); 30m < ht < 1000m (4.68)
G(hr) = 10 log10(hr=3); hr 3m (4.69)
G(hr) = 20 log10(hr=3); 3m < hr < 10m (4.70)
Correlation factors related to terrain are also developed in order to improve the
models accuracy. Okumura's model has a 10-14 dB empirical standard deviation
between the path loss predicted by the model and the path loss associated with one
of the measurements used to develop the model.
4.7.2 Hata Model
The Hata model is an empirical formulation of the graphical path-loss data provided
by the Okumura and is valid over roughly the same range of frequencies, 150-1500
MHz. This empirical formula simpli es the calculation of path loss because it is
closed form formula and it is not based on empirical curves for the di erent parameters.
The standard formula for empirical path loss in urban areas under the Hata
model is
PL;urban(d)dB = 69:55+26:16 log10(fc)􀀀13:82 log10(ht)􀀀a(hr)+(44:9􀀀6:55 log10(ht)) log10(d)
(4.71)
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The parameters in this model are same as in the Okumura model,and a(hr) is a
correction factor for the mobile antenna height based on the size of coverage area.For
small to medium sized cities this factor is given by
a(hr) = (1:11 log10(fc) 􀀀 0:7)hr 􀀀 (1:56 log10(fc) 􀀀 0:8)dB
and for larger cities at a frequencies fc > 300 MHz by
a(hr) = 3:2(log10(11:75hr))2 􀀀 4:97dB
else it is
a(hr) = 8:29(log10(1:54hr))2 􀀀 1:1dB
Corrections to the urban model are made for the suburban, and is given by
PL;suburban(d)dB = PL;urban(d)dB 􀀀 2(log10(fc=28))2 􀀀 5:4 (4.72)
Unlike the Okumura model,the Hata model does not provide for any speci c pathcorrelation
factors. The Hata model well approximates the Okumura model for
distances d > 1 Km. Hence it is a good model for rst generation cellular systems,
but it does not model propogation well in current cellular systems with smaller cell
sizes and higher frequencies. Indoor environments are also not captured by the Hata
model.
4.8 Indoor Propagation Models
The indoor radio channel di ers from the traditional mobile radio channel in ways
- the distances covered are much smaller ,and the variability of the environment
is much greater for smaller range of Tx-Rx separation distances.Features such as
lay-out of the building,the construction materials,and the building type strongly in-

uence the propagation within the building.Indoor radio propagation is dominated
by the same mechanisms as outdoor: re
ection, di raction and scattering with variable
conditions. In general,indoor channels may be classi ed as either line-of-sight
or obstructed.
4.8.1 Partition Losses Inside a Floor (Intra-
oor)
The internal and external structure of a building formed by partitions and obstacles
vary widely.Partitions that are formed as a part of building structure are called
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hard partitions , and partitions that may be moved and which do not span to
the ceiling are called soft partitions. Partitions vary widely in their physical and
electrical characteristics,making it di cult to apply general models to speci c indoor
installations.
4.8.2 Partition Losses Between Floors (Inter-
oor)
The losses between
oors of a building are determined by the external dimensions
and materials of the building,as well as the type of construction used to create the

oors and the external surroundings. Even the number of windows in a building
and the presence of tinting can impact the loss between
oors.
4.8.3 Log-distance Path Loss Model
It has been observed that indoor path loss obeys the distance power law given by
PL(dB) = PL(d0) + 10n log10(d=d0) + X (4.73)
where n depends on the building and surrounding type, and X represents a normal
random variable in dB having standard deviation of dB.
4.9 Summary
In this chapter, three principal propagation models have been identi ed: free-space
propagation, re
ection and di raction, which are common terrestrial models and
these mainly explains the large scale path loss. Regarding path-loss, one important
factor introduced in this chapter is log-distance path loss model. These, however,
may be insigni cant when we consider the small-scale rapid path losses. This is
discussed in the next chapter.
4.10 References
1. T. S. Rappaport, Wireless Communications: Principles and Practice, 2nd ed.
Singapore: Pearson Education, Inc., 2002.
2. S. Haykin and M. Moher, Modern Wireless Communications. Singapore: Pearson
Education, Inc., 2002.
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3. J. W. Mark and W. Zhuang, Wireless Communications and Networking. New
Delhi: PHI, 2005.
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Chapter 5
Multipath Wave Propagation
and Fading
5.1 Multipath Propagation
In wireless telecommunications, multipath is the propagation phenomenon that results
in radio signals reaching the receiving antenna by two or more paths. Causes
of multipath include atmospheric ducting, ionospheric re
ection and refraction, and
re
ection from water bodies and terrestrial objects such as mountains and buildings.
The e ects of multipath include constructive and destructive interference, and phase
shifting of the signal. In digital radio communications (such as GSM) multipath can
cause errors and a ect the quality of communications. We discuss all the related
issues in this chapter.
5.2 Multipath & Small-Scale Fading
Multipath signals are received in a terrestrial environment, i.e., where di erent forms
of propagation are present and the signals arrive at the receiver from transmitter via
a variety of paths. Therefore there would be multipath interference, causing multipath
fading. Adding the e ect of movement of either Tx or Rx or the surrounding
clutter to it, the received overall signal amplitude or phase changes over a small
amount of time. Mainly this causes the fading.
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5.2.1 Fading
The term fading, or, small-scale fading, means rapid
uctuations of the amplitudes,
phases, or multipath delays of a radio signal over a short period or short travel
distance. This might be so severe that large scale radio propagation loss e ects
might be ignored.
5.2.2 Multipath Fading E ects
In principle, the following are the main multipath e ects:
1. Rapid changes in signal strength over a small travel distance or time interval.
2. Random frequency modulation due to varying Doppler shifts on di erent multipath
signals.
3. Time dispersion or echoes caused by multipath propagation delays.
5.2.3 Factors In
uencing Fading
The following physical factors in
uence small-scale fading in the radio propagation
channel:
(1) Multipath propagation { Multipath is the propagation phenomenon that results
in radio signals reaching the receiving antenna by two or more paths.
The e ects of multipath include constructive and destructive interference, and
phase shifting of the signal.
(2) Speed of the mobile { The relative motion between the base station and the
mobile results in random frequency modulation due to di erent doppler shifts
on each of the multipath components.
(3) Speed of surrounding objects { If objects in the radio channel are in motion,
they induce a time varying Doppler shift on multipath components. If
the surrounding objects move at a greater rate than the mobile, then this e ect
dominates fading.
(4) Transmission Bandwidth of the signal { If the transmitted radio signal
bandwidth is greater than the \bandwidth" of the multipath channel (quanti-
ed by coherence bandwidth), the received signal will be distorted.
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5.3 Types of Small-Scale Fading
The type of fading experienced by the signal through a mobile channel depends
on the relation between the signal parameters (bandwidth, symbol period) and the
channel parameters (rms delay spread and Doppler spread). Hence we have four
di erent types of fading. There are two types of fading due to the time dispersive
nature of the channel.
5.3.1 Fading E ects due to Multipath Time Delay Spread
Flat Fading
Such types of fading occurs when the bandwidth of the transmitted signal is less than
the coherence bandwidth of the channel. Equivalently if the symbol period of the
signal is more than the rms delay spread of the channel, then the fading is
at fading.
So we can say that
at fading occurs when
BS BC (5.1)
where BS is the signal bandwidth and BC is the coherence bandwidth. Also
TS (5.2)
where TS is the symbol period and is the rms delay spread. And in such a case,
mobile channel has a constant gain and linear phase response over its bandwidth.
Frequency Selective Fading
Frequency selective fading occurs when the signal bandwidth is more than the coherence
bandwidth of the mobile radio channel or equivalently the symbols duration
of the signal is less than the rms delay spread.
BS BC (5.3)
and
TS (5.4)
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At the receiver, we obtain multiple copies of the transmitted signal, all attenuated
and delayed in time. The channel introduces inter symbol interference. A rule of
thumb for a channel to have
at fading is if

TS
0:1 (5.5)
5.3.2 Fading E ects due to Doppler Spread
Fast Fading
In a fast fading channel, the channel impulse response changes rapidly within the
symbol duration of the signal. Due to Doppler spreading, signal undergoes frequency
dispersion leading to distortion. Therefore a signal undergoes fast fading if
TS TC (5.6)
where TC is the coherence time and
BS BD (5.7)
where BD is the Doppler spread. Transmission involving very low data rates su er
from fast fading.
Slow Fading
In such a channel, the rate of the change of the channel impulse response is much
less than the transmitted signal. We can consider a slow faded channel a channel in
which channel is almost constant over atleast one symbol duration. Hence
TS TC (5.8)
and
BS BD (5.9)
We observe that the velocity of the user plays an important role in deciding whether
the signal experiences fast or slow fading.
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Figure 5.1: Illustration of Doppler e ect.
5.3.3 Doppler Shift
The Doppler e ect (or Doppler shift) is the change in frequency of a wave for an
observer moving relative to the source of the wave. In classical physics (waves in
a medium), the relationship between the observed frequency f and the emitted
frequency fo is given by:
f =

v vr
v vs

f0 (5.10)
where v is the velocity of waves in the medium, vs is the velocity of the source
relative to the medium and vr is the velocity of the receiver relative to the medium.
In mobile communication, the above equation can be slightly changed according
to our convenience since the source (BS) is xed and located at a remote elevated
level from ground. The expected Doppler shift of the EM wave then comes out to
be vr
c fo or, vr
. As the BS is located at an elevated place, a cos factor would
also be multiplied with this. The exact scenario, as given in Figure 5.1, is illustrated
below.
Consider a mobile moving at a constant velocity v, along a path segment length
d between points A and B, while it receives signals from a remote BS source S. The
di erence in path lengths traveled by the wave from source S to the mobile at points
A and B is l = d cos = v t cos , where t is the time required for the mobile
to travel from A to B, and is assumed to be the same at points A and B since the
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source is assumed to be very far away. The phase change in the received signal due
to the di erence in path lengths is therefore
' =
2 l

=
2 v t

cos (5.11)
and hence the apparent change in frequency, or Doppler shift (fd) is
fd =
1
2
:
'
t
= v

: cos : (5.12)
Example 1
An aircraft is heading towards a control tower with 500 kmph, at an elevation of
20 . Communication between aircraft and control tower occurs at 900 MHz. Find
out the expected Doppler shift.
Solution As given here,
v = 500 kmph
the horizontal component of the velocity is
v0 = v cos = 500 cos 20 = 130 m=s
Hence, it can be written that
=
900 106
3 108 =
1
3m
fd =
130
1=3
= 390Hz
If the plane banks suddenly and heads for other direction, the Doppler shift change
will be 390Hz to 􀀀390Hz.
5.3.4 Impulse Response Model of a Multipath Channel
Mobile radio channel may be modeled as a linear lter with time varying impulse
response in continuous time. To show this, consider time variation due to receiver
motion and time varying impulse response h(d; t) and x(t), the transmitted signal.
The received signal y(d; t) at any position d would be
y(d; t) = x(t) h(d; t) =
Z 1
􀀀1
x( ) h(d; t 􀀀 ) d (5.13)
For a causal system: h(d; t) = 0, for t < 0 and for a stable system
R1
􀀀1 jh(d; t)j dt <
1
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Applying causality condition in the above equation, h(d; t 􀀀 ) = 0 for t 􀀀 < 0
) > t, i.e., the integral limits are changed to
y(d; t) =
Z t
􀀀1
x( ) h(d; t 􀀀 ) d :
Since the receiver moves along the ground at a constant velocity v, the position of
the receiver is d = vt, i.e.,
y(vt; t) =
Z t
􀀀1
x( ) h(vt; t 􀀀 ) d :
Since v is a constant, y(vt; t) is just a function of t. Therefore the above equation
can be expressed as
y(t) =
Z t
􀀀1
x( ) h(vt; t 􀀀 ) d = x(t) h(vt; t) = x(t) h(d; t) (5.14)
It is useful to discretize the multipath delay axis of the impulse response into equal
time delay segments called excess delay bins, each bin having a time delay width
equal to ( i+1 􀀀 i) = and i = i for i 2 f0; 1; 2; ::N 􀀀1g, where N represents
the total number of possible equally-spaced multipath components, including the
rst arriving component. The useful frequency span of the model is 2/ . The
model may be used to analyze transmitted RF signals having bandwidth less than
2/ .
If there are N multipaths, maximum excess delay is given by N .
fy(t) = x(t) h(t; i)ji = 0; 1; :::N 􀀀 1g (5.15)
Bandpass channel impulse response model is
x(t) ! h(t; ) = Refhb(t; )ej!ct ! y(t) = Refr(t)ej!ctg (5.16)
Baseband equivalent channel impulse response model is given by
c(t) !
1
2hb(t; ) ! r(t) = c(t)
1
2hb(t; ) (5.17)
Average power is
x2(t) =
1
2jc(t)j2 (5.18)
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The baseband impulse response of a multipath channel can be expressed as
hb(t; ) =
NX􀀀1
i=0
ai(t; ) exp[j(2 fc i(t) + 'i(t; ))] ( 􀀀 i(t)) (5.19)
where ai(t; ) and i(t) are the real amplitudes and excess delays, respectively, of
the ith multipath component at time t. The phase term 2 fc i(t) + 'i(t; ) in the
above equation represents the phase shift due to free space propagation of the ith
multipath component, plus any additional phase shifts which are encountered in the
channel.
If the channel impulse response is wide sense stationary over a small-scale time or
distance interval, then
hb( ) =
NX􀀀1
i=0
ai exp[j i] ( 􀀀 i) (5.20)
For measuring hb( ), we use a probing pulse to approximate (t) i.e.,
p(t) (t 􀀀 ) (5.21)
Power delay pro le is taken by spatial average of jhb(t; )j2 over a local area. The
received power delay pro le in a local area is given by
p( ) kjhb(t; )j2: (5.22)
5.3.5 Relation Between Bandwidth and Received Power
In actual wireless communications, impulse response of a multipath channel is measured
using channel sounding techniques. Let us consider two extreme channel
sounding cases.
Consider a pulsed, transmitted RF signal
x(t) = Refp(t)ej2 fctg (5.23)
where p(t) =
q
4 max
Tbb
for 0 t Tbb and 0 elsewhere. The low pass channel output
is
r(t) =
1
2
NX􀀀1
i=0
ai exp[j i]p(t 􀀀 i)
=
NX􀀀1
i=0
ai exp[j i]:
r
max
Tbb
rect(t 􀀀
Tb
2 􀀀 i):
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Figure 5.2: A generic transmitted pulsed RF signal.
The received power at any time t0 is
jr(t0)j2 =
1
max
Zmax
0
r(t)r (t)dt
=
1
max
Zmax
0
1
4

NX􀀀1
k=0
a2
k(t0)p2(t 􀀀 k)
!
dt
=
1
max
NX􀀀1
k=0
a2
k(t0)
Zmax
0
r
max
Tbb
rect(t 􀀀
Tb
2 􀀀 i)
2
dt
=
NX􀀀1
k=0
a2
k(t0):
Interpretation: If the transmitted signal is able to resolve the multipaths, then
average small-scale receiver power is simply sum of average powers received from
each multipath components.
Ea; [PWB] = Ea; [
NX􀀀1
i=0
jai exp(j i)j2]
NX􀀀1
i=0
a2
i (5.24)
Now instead of a pulse, consider a CW signal, transmitted into the same channel
and for simplicity, let the envelope be c(t) = 2. Then
r(t) =
NX􀀀1
i=0
ai exp[j i(t; )] (5.25)
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and the instantaneous power is
jr(t)j2 = j
NX􀀀1
i=0
ai exp[j i(t; )]j2 (5.26)
Over local areas, ai varies little but i varies greatly resulting in large
uctuations.
Ea; [PCW] = Ea; [
NX􀀀1
i=0
jai exp(j i)j2]

NX􀀀1
i=0
a2
i + 2
NX􀀀1
i=0
XN
i;j6=i
rijcos( i 􀀀 j)
where rij = Ea[aiaj ].
If, rij = cos( i 􀀀 j) = 0, then Ea; [PCW] = Ea; [PWB]. This occurs if multipath
components are uncorrelated or if multipath phases are i.i.d over [0; 2 ].
Bottomline:
1. If the signal bandwidth is greater than multipath channel bandwidth then
fading e ects are negligible
2. If the signal bandwidth is less than the multipath channel bandwidth, large
fading occurs due to phase shift of unresolved paths.
5.3.6 Linear Time Varying Channels (LTV)
The time variant transfer function(TF) of an LTV channel is FT of h(t; ) w.r.t. .
H(f; t) = FT[h( ; t)] =
Z 1
􀀀1
h( ; t)e􀀀j2 f d (5.27)
h( ; t) = FT􀀀1[H(f; t)] =
Z 1
􀀀1
H(f; t)ej2 f df (5.28)
The received signal
r(t) =
Z 1
􀀀1
R(f; t)ej2 ft df (5.29)
where R(f; t) = H(f; t)X(f).
For
at fading channel, h( ; t) = Z(t) ( 􀀀 i) where Z(t) =
P
n(t)e􀀀j2 fc n(t). In
this case, the received signal is
r(t) =
Z 1
􀀀1
h( ; t)x(t 􀀀 ) d = Z(t)x(t 􀀀 i) (5.30)
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Figure 5.3: Relationship among di erent channel functions.
where the channel becomes multiplicative.
Doppler spread functions:
H(f; ) = FT[H(f; t)] =
Z 1
􀀀1
H(f; t)e􀀀j2 tdt (5.31)
and
H(f; t) = FT􀀀1[H(f; )] =
Z 1
􀀀1
H(f; )ej2 td (5.32)
Delay Doppler spread:
H( ; ) = FT[h( ; t)] =
Z 1
􀀀1
h( ; t)e􀀀j2 tdt (5.33)
5.3.7 Small-Scale Multipath Measurements
Direct RF Pulse System
A wideband pulsed bistatic radar usually transmits a repetitive pulse of width Tbb s,
and uses a receiver with a wide bandpass lter (BW = 2
Tbb
Hz). The signal is then
ampli ed, envelope detected, and displayed and stored on a high speed oscilloscope.
Immediate measurements of the square of the channel impulse response convolved
with the probing pulse can be taken. If the oscilloscope is set on averaging mode,
then this system provides a local average power delay pro le.
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Figure 5.4: Direct RF pulsed channel IR measurement.
This system is subject to interference noise. If the rst arriving signal is blocked
or fades, severe fading occurs, and it is possible the system may not trigger properly.
Frequency Domain Channel Sounding
In this case we measure the channel in the frequency domain and then convert it
into time domain impulse response by taking its inverse discrete Fourier transform
(IDFT). A vector network analyzer controls a swept frequency synthesizer. An Sparameter
test set is used to monitor the frequency response of the channel. The
sweeper scans a particular frequency band, centered on the carrier, by stepping
through discrete frequencies. The number and spacing of the frequency step impacts
the time resolution of the impulse response measurement. For each frequency step,
the S-parameter test set transmits a known signal level at port 1 and monitors
the received signal at port 2. These signals allow the analyzer to measure the
complex response, S21(!), of the channel over the measured frequency range. The
S21(!) measure is the measure of the signal
ow from transmitter antenna to receiver
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Figure 5.5: Frequency domain channel IR measurement.
antenna (i.e., the channel).
This system is suitable only for indoor channel measurements. This system is
also non real-time. Hence, it is not suitable for time-varying channels unless the
sweep times are fast enough.
5.4 Multipath Channel Parameters
To compare the di erent multipath channels and to quantify them, we de ne some
parameters. They all can be determined from the power delay pro le. These parameters
can be broadly divided in to two types.
5.4.1 Time Dispersion Parameters
These parameters include the mean excess delay,rms delay spread and excess delay
spread. The mean excess delay is the rst moment of the power delay pro le and is
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de ned as
=
P
a2
P k k
a2
k
=
P
PP( k) k
P( k)
(5.34)
where ak is the amplitude, k is the excess delay and P( k) is the power of the
individual multipath signals.
The mean square excess delay spread is de ned as
2 =
P
P( k) 2
P k
P( k)
(5.35)
Since the rms delay spread is the square root of the second central moment of the
power delay pro le, it can be written as
=
q
2 􀀀 ( )2 (5.36)
As a rule of thumb, for a channel to be
at fading the following condition must be
satis ed

TS
0:1 (5.37)
where TS is the symbol duration. For this case, no equalizer is required at the
receiver.
Example 2
1. Sketch the power delay pro le and compute RMS delay spread for the following:
P( ) =
P1
n=0
( 􀀀 n 10􀀀6) (in watts)
2. If BPSK modulation is used, what is the maximum bit rate that can be sent
through the channel without needing an equalizer?
Solution
1. P(0) = 1 watt, P(1) = 1 watt
=
(1)(0) + (1)(1)
1 + 1
= 0:5 s
2 = 0:5 s2 = 0:5 s
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2. For
at fading channel, we need
Ts
0:1 ) Rs = 1
Ts
= 0:2 104 = 200 kbps
For BPSK we need Rb = Rs = 200 kbps
Example 3 A simple delay spread bound: Feher's upper bound
Consider a simple worst-case delay spread scenario as shown in gure below.
Here dmin = d0 and dmax = di + dr
Transmitted power = PT , Minimum received power = PRmin = PThreshold
PRmin
PT
= GTGR(
4 dmax
)2
Put GT = GR = 1 i.e., considering omni-directional unity gain antennas
dmax = (
4
)( PT
PRmin
) 1
2
max = dmax
c
= (
4 c
)( PT
PRmin
) 1
2
max = (
1
4 f
)( PT
PRmin
) 1
2
5.4.2 Frequency Dispersion Parameters
To characterize the channel in the frequency domain, we have the following parameters.
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(1) Coherence bandwidth: it is a statistical measure of the range of frequencies
over which the channel can be considered to pass all the frequency components with
almost equal gain and linear phase. When this condition is satis ed then we say the
channel to be
at.
Practically, coherence bandwidth is the minimum separation over which the two
frequency components are a ected di erently. If the coherence bandwidth is considered
to be the bandwidth over which the frequency correlation function is above
0.9, then it is approximated as
BC
1
50
: (5.38)
However, if the coherence bandwidth is considered to be the bandwidth over which
the frequency correlation function is above 0.5, then it is de ned as
BC
1
5
: (5.39)
The coherence bandwidth describes the time dispersive nature of the channel in the
local area. A more convenient parameter to study the time variation of the channel
is the coherence time. This variation may be due to the relative motion between the
mobile and the base station or the motion of the objects in the channel.
(2) Coherence time: this is a statistical measure of the time duration over which
the channel impulse response is almost invariant. When channel behaves like this,
it is said to be slow faded. Essentially it is the minimum time duration over which
two received signals are a ected di erently. For an example, if the coherence time
is considered to be the bandwidth over which the time correlation is above 0.5, then
it can be approximated as
TC
9
16 fm
(5.40)
where fm is the maximum doppler spread given be fm =
.
Another parameter is the Doppler spread (BD) which is the range of frequencies
over which the received Doppler spectrum is non zero.
5.5 Statistical models for multipath propagation
Many multipath models have been proposed to explain the observed statistical nature
of a practical mobile channel. Both the rst order and second order statistics
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Figure 5.6: Two ray NLoS multipath, resulting in Rayleigh fading.
have been examined in order to nd out the e ective way to model and combat the
channel e ects. The most popular of these models are Rayleigh model, which describes
the NLoS propagation. The Rayleigh model is used to model the statistical
time varying nature of the received envelope of a
at fading envelope. Below, we
discuss about the main rst order and second order statistical models.
5.5.1 NLoS Propagation: Rayleigh Fading Model
Let there be two multipath signals S1 and S2 received at two di erent time instants
due to the presence of obstacles as shown in Figure 5.6. Now there can either be
constructive or destructive interference between the two signals.
Let En be the electric eld and n be the relative phase of the various multipath
signals.So we have
~E
=
XN
n=1
Enej n (5.41)
Now if N! 1(i.e. are su ciently large number of multipaths) and all the En are
IID distributed, then by Central Limit Theorem we have,
lim
N!1
~E
= lim
N!1
XN
n=1
Enej n (5.42)
= Zr + jZi = Rej (5.43)
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where Zr and Zi are Gaussian Random variables. For the above case
R =
q
Z2
r + Z2
i (5.44)
and
= tan􀀀1 Zi
Zr
(5.45)
For all practical purposes we assume that the relative phase n is uniformaly distributed.
E[ej n] =
1
2
Z2
0
ej d = 0 (5.46)
It can be seen that En and n are independent. So,
E[~E ] = E[
X
Enej n] = 0 (5.47)
E[

~E

2] = E[
X
Enej n
X
E
ne􀀀j n] = E[
X
m
X
n
EnEmej( n􀀀 m)] =
XN
n=1
E2
n = P0
(5.48)
where P0 is the total power obtained. To nd the Cumulative Distribution Function(
CDF) of R, we proceed as follows.
FR(r) = Pr(R r) =
Z
A
Z
fZi;Zr (zi; zr)dzidzr (5.49)
where A is determined by the values taken by the dummy variable r. Let Zi and Zr
be zero mean Gaussian RVs. Hence the CDF can be written as
FR(r) =
Z
A
Z 1
p
2 2
e
􀀀(Z2
r+Z2
i
)
2 2 dZidZr (5.50)
Let Zr = p cos( ) and Zi = p sin( ) So we have
FR(r) =
Z2
0
Z2
0
1
p
2 2
e
􀀀p2
2 2 pdpd (5.51)
= 1 􀀀 e
􀀀r2
2 2 (5.52)
Above equation is valid for all r 0. The pdf can be written as
fR(r) = r
2 e􀀀 r2
2 2 (5.53)
and is shown in Figure 5.7 with di erent values. This equation too is valid for all
r 0. Above distribution is known as Rayleigh distribution and it has been derived
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Figure 5.7: Rayleigh probability density function.
for slow fading. However, if fD 1 Hz, we call it as Quasi-stationary Rayleigh
fading. We observe the following:
E[R] =
r

2 (5.54)
E[R2] = 2 2 (5.55)
var[R] = (2 􀀀

2
) 2 (5.56)
median[R] = 1:77 : (5.57)
5.5.2 LoS Propagation: Rician Fading Model
Rician Fading is the addition to all the normal multipaths a direct LOS path.
Figure 5.8: Ricean probability density function.
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fR(r) = r
2 e
􀀀(r2+A2)
2 2 I0(Ar
2 ) (5.58)
for all A 0 and r 0. Here A is the peak amplitude of the dominant signal and
I0(:) is the modi ed Bessel function of the rst kind and zeroth order.
A factor K is de ned as
KdB = 10 log A2
2 2 (5.59)
As A ! 0 then KdB ! 1.
5.5.3 Generalized Model: Nakagami Distribution
A generalization of the Rayleigh and Rician fading is the Nakagami distribution.
Figure 5.9: Nakagami probability density function.
Its pdf is given as,
fR(r) =
2rm􀀀1
􀀀(m)
(mm

m )e
􀀀mr2

 (5.60)
where,
􀀀(m) is the gamma function

 is the average signal power and
m is the fading factor.It is always greater than or equal to 0.5.
When m=1, Nakagami model is the Rayleigh model.
When
m =
(M + 1)2
2M + 1
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Figure 5.10: Schematic representation of level crossing with a Rayleigh fading envelope
at 10 Hz Doppler spread.
where
M = A
2
Nakagami fading is the Rician fading.
As m ! 1 Nakagami fading is the impulse channel and no fading occurs.
5.5.4 Second Order Statistics
To design better error control codes, we have two important second order parameters
of fading model, namely the level crossing rate (LCR) and average fade
duration (AFD). These parameters can be utilized to assess the speed of the user
by measuring them through the reverse channel. The LCR is the expected rate at
which the Rayleigh fading envelope normalized to the local rms amplitude crosses a
speci c level 'R' in a positive going direction.
NR =
1 Z
0
r_p(R; r_)dr_ =
p
2 fD e􀀀 2 (5.61)
where r_ is the time derivative of r(t), fD is the maximum Doppler shift and is the
value of the speci ed level R, normalized to the local rms amplitude of the fading
envelope.
The other important parameter, AFD, is the average period time for which the
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receiver power is below a speci ed level R.
=
1
Nr
Pr(r R) (5.62)
As
Pr(r R) =
ZR
0
p(r)dr = 1 􀀀 e􀀀 2
; (5.63)
therefore,
=
1 􀀀 e􀀀 2
p
2 fD e􀀀 2 (5.64)
= e􀀀 2
􀀀 1
p
2 fD
: (5.65)
Apart from LCR, another parameter is fading rate, which is de ned as the number of
times the signal envelope crosses the middle value (rm) in a positive going direction
per unit time. The average rate is expressed as
N(rm) =
2v

: (5.66)
Another statistical parameter, sometimes used in the mobile communication, is
called as depth of fading. It is de ned as the ratio between the minimum value
and the mean square value of the faded signal. Usually, an average value of 10% as
depth of fading gives a marginal fading scenario.
5.6 Simulation of Rayleigh Fading Models
5.6.1 Clarke's Model: without Doppler E ect
In it, two independent Gaussian low pass noise sources are used to produce in-phase
and quadrature fading branches. This is the basic model and is useful for slow fading
channel. Also the Doppler e ect is not accounted for.
5.6.2 Clarke and Gans' Model: with Doppler E ect
In this model, the output of the Clarke's model is passed through Doppler lter in
the RF or through two initial baseband Doppler lters for baseband processing as
shown in Figure 5.11. Here, the obtained Rayleigh output is
at faded signal but
not frequency selective.
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Figure 5.11: Clarke and Gan's model for Rayleigh fading generation using quadrature
amplitude modulation with (a) RF Doppler lter, and, (b) baseband Doppler
lter.
5.6.3 Rayleigh Simulator with Wide Range of Channel Conditions
To get a frequency selective output we have the following simulator through which
both the frequency selective and
at faded Rayleigh signal may be obtained. This
is achieved through varying the parameters ai and i, as given in Figure 5.12.
5.6.4 Two-Ray Rayleigh Faded Model
The above model is, however, very complex and di cult to implement. So, we have
the two ray Rayleigh fading model which can be easily implemented in software as
shown in Figure 5.13.
hb(t) = 1ej 1 (t) + 2ej 2 (t 􀀀 ) (5.67)
where 1 and 2 are independent Rayleigh distributed and 1 and 2 are independent
and uniformaly distributed over 0 to 2 . By varying it is possible to create
a wide range of frequency selective fading e ects.
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Figure 5.12: Rayleigh fading model to get both the
at and frequency selective
channel conditions.
5.6.5 Saleh and Valenzuela Indoor Statistical Model
This method involved averaging the square law detected pulse response while sweeping
the frequency of the transmitted pulse. The model assumes that the multipath
components arrive in clusters. The amplitudes of the received components are independent
Rayleigh random variables with variances that decay exponentially with
cluster delay as well as excess delay within a cluster. The clusters and multipath
components within a cluster form Poisson arrival processes with di erent rates.
5.6.6 SIRCIM/SMRCIM Indoor/Outdoor Statistical Models
SIRCIM (Simulation of Indoor Radio Channel Impulse-response Model) generates
realistic samples of small-scale indoor channel impulse response measurements. Sub-
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Figure 5.13: Two-ray Rayleigh fading model.
sequent work by Huang produced SMRCIM (Simulation of Mobile Radio Channel
Impulse-response Model), a similar program that generates small-scale urban cellular
and micro-cellular channel impulse responses.
5.7 Conclusion
In this chapter, the main channel impairment, i.e., fading, has been introduced which
becomes so severe sometimes that even the large scale path loss becomes insigni cant
in comparison to it. Some statistical propagation models have been presented based
on the fading characteristics. Mainly the frequency selective fading, fast fading and
deep fading can be considered the major obstruction from the channel severity view
point. Certain e cient signal processing techniques to mitigate these e ects will be
discussed in Chapter 7.
5.8 References
1. T. S. Rappaport, Wireless Communications: Principles and Practice, 2nd ed.
Singapore: Pearson Education, Inc., 2002.
2. S. Haykin and M. Moher, Modern Wireless Communications. Singapore: Pearson
Education, Inc., 2002.
3. J. W. Mark and W. Zhuang, Wireless Communications and Networking. New
Delhi: PHI, 2005.
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4. K. Feher, Wireless Digital Communications: Modulation and Spread Spectrum
Applications. Upper Saddle River, NJ: Prentice Hall, 1995.
5. R. Blake, Wireless Communications Technology. Delmar, Singapore: Thomson
Asia Pvt Ltd, 2004.
6. D. P. Agarwal and Q-A. Zeng, Introduction to Wireless and Mobile Systems.
Nelson, India: Thomson Learning, 2007.
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Chapter 6
Transmitter and Receiver
Techniques
6.1 Introduction
Electrical communication transmitter and receiver techniques strive toward obtaining
reliable communication at a low cost, with maximum utilization of the channel
resources. The information transmitted by the source is received by the destination
via a physical medium called a channel. This physical medium, which may be
wired or wireless, introduces distortion, noise and interference in the transmitted
information bearing signal. To counteract these e ects is one of the requirements
while designing a transmitter and receiver end technique. The other requirements
are power and bandwidth e ciency at a low implementation complexity.
6.2 Modulation
Modulation is a process of encoding information from a message source in a manner
suitable for transmission. It involves translating a baseband message signal to
a passband signal. The baseband signal is called the modulating signal and the
passband signal is called the modulated signal. Modulation can be done by varying
certain characteristics of carrier waves according to the message signal. Demodulation
is the reciprocal process of modulation which involves extraction of original
baseband signal from the modulated passband signal.
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6.2.1 Choice of Modulation Scheme
Several factors in
uence the choice of a digital modulation scheme. A desirable
modulation scheme provides low bit error rates at low received signal to noise ratios,
performs well in multipath and fading conditions, occupies a minimum of bandwidth,
and is easy and cost-e ective to implement. The performance of a modulation
scheme is often measured in terms of its power e ciency and bandwidth e ciency.
Power e ciency describes the ability of a modulation technique to preserve the
delity of the digital message at low power levels. In a digital communication system,
in order to increase noise immunity, it is necessary to increase the signal power.
Bandwidth e ciency describes the ability of a modulation scheme to accommodate
data within a limited bandwidth.
The system capacity of a digital mobile communication system is directly related
to the bandwidth e ciency of the modulation scheme, since a modulation with a
greater value of b(= R
B) will transmit more data in a given spectrum allocation.
There is a fundamental upper bound on achievable bandwidth e ciency. Shannon's
channel coding theorem states that for an arbitrarily small probability of error,
the maximum possible bandwidth e ciency is limited by the noise in the channel,
and is given by the channel capacity formula
Bmax = C
B
= log2(1 + S
N
) (6.1)
6.2.2 Advantages of Modulation
1. Facilitates multiple access: By translating the baseband spectrum of signals
from various users to di erent frequency bands, multiple users can be accommodated
within a band of the electromagnetic spectrum.
2. Increases the range of communication: Low frequency baseband signals su er
from attenuation and hence cannot be transmitted over long distances. So
translation to a higher frequency band results in long distance transmission.
3. Reduction in antenna size: The antenna height and aperture is inversely proportional
to the radiated signal frequency and hence high frequency signal
radiation result in smaller antenna size.
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6.2.3 Linear and Non-linear Modulation Techniques
The mathematical relation between the message signal (applied at the modulator
input) and the modulated signal (obtained at the modulator output) decides whether
a modulation technique can be classi ed as linear or non-linear. If this input-output
relation satis es the principle of homogeneity and superposition then the modulation
technique is said to be linear. The principle of homogeneity states that if the input
signal to a system (in our case the system is a modulator) is scaled by a factor then
the output must be scaled by the same factor. The principle of superposition states
that the output of a linear system due to many simultaneously applied input signals
is equal to the summation of outputs obtained when each input is applied one at a
time.
For example an amplitude modulated wave consists of the addition two terms: the
message signal multiplied with the carrier and the carrier itself. If m(t) is the
message signal and sAM(t) is the modulated signal given by:
sAM(t) = Ac[1 + km(t)] cos(2 fct) (6.2)
Then,
1. From the principle of homogeneity: Let us scale the input by a factor a. So
m(t) = am1(t) and the corresponding output becomes :
sAM1(t) = Ac[1 + am1(t)] cos(2 fct) (6.3)
6= asAM1(t)
2. From the principle of superposition: Let m(t) = m1(t) + m2(t) be applied
simultaneously at the input of the modulator. The resulting output is:
sAM(t) = Ac[1 + m1(t) + m2(t)] cos(2 fct) (6.4)
6= sAM1(t) + sAM2(t)
= Ac[2 + m1(t) + m2(t)] cos(2 fct)
Here, sAM1(t) and sAM2(t) are the outputs obtained when m1(t) and m2(t)
are applied one at a time.
Hence AM is a nonlinear technique but DSBSC modulation is a linear technique
since it satis es both the above mentioned principles.
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6.2.4 Amplitude and Angle Modulation
Depending on the parameter of the carrier (amplitude or angle) that is changed
in accordance with the message signal, a modulation scheme can be classi ed as
an amplitude or angle modulation. Amplitude modulation involves variation of
amplitude of the carrier wave with changes in the message signal. Angle modulation
varies a sinusoidal carrier signal in such a way that the angle of the carrier is varied
according to the amplitude of the modulating baseband signal.
6.2.5 Analog and Digital Modulation Techniques
The nature of the information generating source classi es a modulation technique as
an analog or digital modulation technique. When analog messages generated from
a source passe through a modulator, the resulting amplitude or angle modulation
technique is called analog modulation. When digital messages undergo modulation
the resulting modulation technique is called digital modulation.
6.3 Signal Space Representation of Digitally Modulated
Signals
Any arbitrary signal can be expressed as the linear combination of a set of orthogonal
signals or equivalently as a point in an M dimensional signal space, where M
denotes the cardinality of the set of orthogonal signals. These orthogonal signals are
normalized with respect to their energy content to yield an orthonormal signal set
having unit energy. These orthonormal signals are independent of each other and
form a basis set of the signal space.
Generally a digitally modulated signal s(t), having a symbol duration T, is expressed
as a linear combination of two orthonormal signals 1(t) and 2(t), constituting
the two orthogonal axis in this two dimensional signal space and is expressed
mathematically as,
s(t) = s1 1(t) + s2 2(t) (6.5)
where 1(t) and 2(t) are given by,
1(t) =
r
2
T
cos(2 fct) (6.6)
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2(t) =
r
2
T
cos(2 fct) (6.7)
The coe cients s1 and s2 form the coordinates of the signal s(t) in the two dimensional
signal space.
6.4 Complex Representation of Linear Modulated Sig-
nals and Band Pass Systems
A band-pass signal s(t) can be resolved in terms of two sinusoids in phase quadrature
as follows:
s(t) = sI (t)cos(2 fct) 􀀀 sQ(t)sin(2 fct) (6.8)
Hence sI (t) and sQ(t) are known as the in-phase and quadrature-phase components
respectively. When sI (t) and sQ(t) are incorporated in the formation of the following
complex signal,
~s(t) = sI (t) + sQ(t) (6.9)
then s(t) can be expressed in a more compact form as:
s(t) = Ref~s(t)e(j2 fct)g (6.10)
where ~s(t) is called the complex envelope of s(t).
Analogously, band-pass systems characterized by an impulse response h(t) can
be expressed in terms of its in-phase and quadrature-phase components as:
h(t) = hI (t)cos(2 fct) 􀀀 hQ(t)sin(2 fct) (6.11)
The complex baseband model for the impulse response therefore becomes,
~h(
t
)
=
h
I
(
t
)
+
h
Q
(
t
)
(
6.12)
h(t) can therefore be expressed in terms of its complex envelope as
h(t) = Ref~h(t)ej2 fctg: (6.13)
When s(t) passes through h(t), then in the complex baseband domain, the output
~r(t) of the bandpass system is given by the following convolution
~r(t) =
1
2
~s(t)
~h(t) (6.14)
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6.5 Linear Modulation Techniques
6.5.1 Amplitude Modulation (DSBSC)
Generally, in amplitude modulation, the amplitude of a high frequency carrier signal,
cos(2 fct), is varied in accordance to the instantaneous amplitude of the modulating
message signal m(t). The resulting modulated carrier or AM signal can be
represented as:
sAM(t) = Ac[1 + km(t)] cos(2 fct): (6.15)
The modulation index k of an AM signal is de ned as the ratio of the peak message
signal amplitude to the peak carrier amplitude. For a sinusoidal modulating signal
m(t) = Am
Ac
cos(2 fmt), the modulation index is given by
k = Am
Ac
: (6.16)
This is a nonlinear technique and can be made linear by multiplying the carrier with
the message signal.The resulting modulation scheme is known as DSBSC modulation.
In DSBSC the amplitude of the transmitted signal, s(t), varies linearly with
the modulating digital signal, m(t). Linear modulation techniques are bandwidth
e cient and hence are very attractive for use in wireless communication systems
where there is an increasing demand to accommodate more and more users within
a limited spectrum. The transmitted signal DSBSC signal s(t) can be expressed as:
s(t) = Am(t)exp(j2 fct): (6.17)
If m(t) is scaled by a factor of a, then s(t), the output of the modulator, is also
scaled by the same factor as seen from the above equation. Hence the principle of
homogeneity is satis ed. Moreover,
s12(t) = A[m1(t) + m2(t)]cos(2 fct) (6.18)
= Am1(t)cos(2 fct) + Am2(t)cos(2 fct)
= s1(t) + s2(t)
where A is the carrier amplitude and fc is the carrier frequency. Hence the principle
of superposition is also satis ed. Thus DSBSC is a linear modulation technique.
AM demodulation techniques may be broadly divided into two categories: coherent
and non-coherent demodulation. Coherent demodulation requires knowledge
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Figure 6.1: BPSK signal constellation.
of the transmitted carrier frequency and phase at the receiver, whereas non-coherent
detection requires no phase information.
6.5.2 BPSK
In binary phase shift keying (BPSK), the phase of a constant amplitude carrier
signal is switched between two values according to the two possible signals m1 and
m2 corresponding to binary 1 and 0, respectively. Normally, the two phases are
separated by 180o. If the sinusoidal carrier has an amplitude A, and energy per bit
Eo = 1
2A2
c Tb then the transmitted BPSK signal is
sBPSK(t) = m(t)
s
2Eb
Tb
cos(2 fct + c): (6.19)
A typical BPSK signal constellation diagram is shown in Figure 6.1.
The probability of bit error for many modulation schemes in an AWGN channel
is found using the Q-function of the distance between the signal points. In case of
BPSK,
PeBPSK = Q(
s
2Eb
N0
): (6.20)
6.5.3 QPSK
The Quadrature Phase Shift Keying (QPSK) is a 4-ary PSK signal. The phase of
the carrier in the QPSK takes 1 of 4 equally spaced shifts. Although QPSK can
be viewed as a quaternary modulation, it is easier to see it as two independently
modulated quadrature carriers. With this interpretation, the even (or odd) bits are
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Figure 6.2: QPSK signal constellation.
Figure 6.3: QPSK transmitter.
used to modulate the in-phase component of the carrier, while the odd (or even)
bits are used to modulate the quadrature-phase component of the carrier.
The QPSK transmitted signal is de ned by:
si(t) = Acos(!t + (i 􀀀 1) =2); i = (1; 2; 3; 4) (6.21)
and the constellation disgram is shown in Figure 6.2.
6.5.4 O set-QPSK
As in QPSK, as shown in Figure 6.3, the NRZ data is split into two streams of odd
and even bits. Each bit in these streams has a duration of twice the bit duration,
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Figure 6.4: DQPSK constellation diagram.
Tb, of the original data stream. These odd (d1(t)) and even bit streams (d2(t)) are
then used to modulate two sinusoidals in phase quadrature,and hence these data
streams are also called the in-phase and and quadrature phase components. After
modulation they are added up and transmitted. The constellation diagram of O set-
QPSK is the same as QPSK. O set-QPSK di ers from QPSK in that the d1(t) and
d2(t) are aligned such that the timing of the pulse streams are o set with respect
to each other by Tb seconds. From the constellation diagram it is observed that a
signal point in any quadrant can take a value in the diagonally opposite quadrant
only when two pulses change their polarities together leading to an abrupt 180 degree
phase shift between adjacent symbol slots. This is prevented in O-QPSK and the
allowed phase transitions are 90 degree.
Abrupt phase changes leading to sudden changes in the signal amplitude in the
time domain corresponds to signi cant out of band high frequency components in
the frequency domain. Thus to reduce these sidelobes spectral shaping is done at
baseband. When high e ciency power ampli ers, whose non-linearity increases as
the e ciency goes high, are used then due to distortion, harmonics are generated
and this leads to what is known as spectral regrowth. Since sudden 180 degree phase
changes cannot occur in OQPSK, this problem is reduced to a certain extent.
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6.5.5 =4 DQPSK
The data for =4 DQPSK like QPSK can be thought to be carried in the phase of a
single modulated carrier or on the amplitudes of a pair of quadrature carriers. The
modulated signal during the time slot of kT < t < (k + 1)T given by:
s(t) = cos(2 fct +  k+1) (6.22)
Here,  k+1 =  k + k and k can take values =4 for 00, 3 =4 for 01, 􀀀3 =4
for 11 and 􀀀 =4 for 10. This corresponds to eight points in the signal constellation
but at any instant of time only one of the four points are possible: the four points
on axis or the four points o axis. The constellation diagram along with possible
transitions are shown in Figure 6.4.
6.6 Line Coding
Speci c waveforms are required to represent a zero and a one uniquely so that a
sequence of bits is coded into electrical pulses. This is known as line coding. There
are various ways to accomplish this and the di erent forms are summarized below.
1. Non-return to zero level (NRZ-L): 1 forces a a high while 0 forces a low.
2. Non-return to zero mark (NRZ-M): 1 forces negative and positive transitions
while 0 causes no transitions.
3. Non-return to zero space (NRZ-S): 0 forces negative and positive transitions
while 1 causes no transitions.
4. Return to zero (RZ): 1 goes high for half a period while 0 remains at zero
state.
5. Biphase-L: Manchester 1 forces positive transition while 0 forces negative transition.
In case of consecutive bits of same type a transition occurs in the
beginning of the bit period.
6. Biphase-M: There is always a transition in the beginning of a bit interval. 1
forces a transition in the middle of the bit while 0 does nothing.
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Figure 6.5: Scematic of the line coding techniques.
7. Biphase-S: There is always a transition in the beginning of a bit interval. 0
forces a transition in the middle of the bit while 1 does nothing.
8. Di erential Manchester: There is always a transition in the middle of a bit
interval. 0 forces a transition in the beginning of the bit while 1 does nothing.
9. Bipolar/Alternate mark inversion (AMI): 1 forces a positive or negative pulse
for half a bit period and they alternate while 0 does nothing.
All these schemes are shown in Figure 6.5.
6.7 Pulse Shaping
Let us think about a rectangular pulse as de ned in BPSK. Such a pulse is not
desirable for two fundamental reasons:
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Figure 6.6: Rectangular Pulse
(a) the spectrum of a rectangular pulse is in nite in extent. Correspondingly, its
frequency content is also in nite. But a wireless channel is bandlimited, means it
would introduce signal distortion to such type of pulses,
(b) a wireless channel has memory due to multipath and therefore it introduces ISI.
In order to mitigate the above two e ects, an e cient pulse shaping funtion or
a premodulation lter is used at the Tx side so that QoS can be maintained to the
mobile users during communication. This type of technique is called pulse shaping
technique. Below, we start with the fundamental works of Nyquist on pulse shaping
and subsequently, we would look into another type of pulse shaping technique.
6.7.1 Nyquist pulse shaping
There are a number of well known pulse shaping techniques which are used to simultaneously
to reduce the inter-symbol e ects and the spectral width of a modulated
digital signal. We discuss here about the fundamental works of Nyquist. As pulse
shaping is di cult to directly manipulate the transmitter spectrum at RF frequencies,
spectral shaping is usually done through baseband or IF processing.
Let the overall frequency response of a communication system (the transmitter,
channel and receiver) be denoted as Heff (f) and according to Nyquist it must be
given by:
Heff (f) =
1
fs
rect( f
fs
) (6.23)
Hence, the ideal pulse shape for zero ISI, given by heff (t), such that,
Heff (f) $ heff (t) (6.24)
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Figure 6.7: Raised Cosine Pulse.
is given by:
heff (t) =
sin( t
Ts
)
t
Ts
(6.25)
(6.26)
6.7.2 Raised Cosine Roll-O Filtering
If we take a rectangular lter with bandwidth f0 1
2Ts
and convolve it with any
arbitrary even function Z(f) with zero magnitude outside the passband of the rectangular
lter then a zero ISI e ect would be achieved. Mathematically,
Heff (f) = rect( f
f0
) Z(f); (6.27)
heff (t) =
sin( t
Ts
)
t
Ts
z(t); (6.28)
z(t) =
cos( t=Ts)
1 􀀀 ( t=2Ts)2 : (6.29)
with being the roll o factor 2 [0; 1]. As increases roll o in frequency domain
increases but that in time domain decreases.
6.7.3 Realization of Pulse Shaping Filters
Since heff (t) is non-causal, pulse shaping lters are usually truncated within 6Ts
about t = 0 for each symbol. Digital communication systems thus often store several
symbols at a time inside the modulator and then clock out a group of symbols by
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using a look up table that represents discrete time waveforms of stored symbols.
This is the way to realize the pulse shaping lters using real time processors.
Non-Nyquist pulse shaping are also useful, which would be discussed later in this
chapter while discussing GMSK.
6.8 Nonlinear Modulation Techniques
Many practical mobile radio communications use nonlinear modulation methods,where
the amplitude of the carrier is constant,regardless of the variations in the modulating
signal.The Constant envelope family of modulations has the following advantages :
1. Power e cient class C ampli ers without introducing degradation in the spectral
occupancy of the transmitted signal.
2. Low out-of-band radiation of the order of -60 dB to -70dB can be achieved.
3. Limiter-discriminator detection can be used,which simpli es receiver design
and provides high immunity against random FM noise and signal
uctuations
due to Rayleigh fading.
However, even if constant envelope has many advantages it still uses more BW
than linear modulation schemes.
6.8.1 Angle Modulation (FM and PM)
There are a number of ways in which the phase of a carrier signal may be varied
in accordance with the baseband signal; the two most important classes of angle
modulation being frequency modulation and phase modulation.
Frequency modulation (FM) involves changing of the frequency of the carrier
signal according to message signal. As the information in frequency modulation is
in the frequency of modulated signal, it is a nonlinear modulation technique. In this
method, the amplitude of the carrier wave is kept constant (this is why FM is called
constant envelope). FM is thus part of a more general class of modulation known
as angle modulation.
Frequency modulated signals have better noise immunity and give better performance
in fading scenario as compared to amplitude modulation.Unlike AM, in an
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FM system, the modulation index, and hence bandwidth occupancy, can be varied
to obtain greater signal to noise performance.This ability of an FM system to trade
bandwidth for SNR is perhaps the most important reason for its superiority over
AM. However, AM signals are able to occupy less bandwidth as compared to FM
signals, since the transmission system is linear.
An FM signal is a constant envelope signal, due to the fact that the envelope of
the carrier does not change with changes in the modulating signal. The constant
envelope of the transmitted signal allows e cient Class C power ampli ers to be
used for RF power ampli cation of FM. In AM, however, it is critical to maintain
linearity between the applied message and the amplitude of the transmitted signal,
thus linear Class A or AB ampli ers, which are not as power e cient, must be used.
FM systems require a wider frequency band in the transmitting media (generally
several times as large as that needed for AM) in order to obtain the advantages of
reduced noise and capture e ect. FM transmitter and receiver equipment is also
more complex than that used by amplitude modulation systems. Although frequency
modulation systems are tolerant to certain types of signal and circuit nonlinearities,
special attention must be given to phase characteristics. Both AM and FM may be
demodulated using inexpensive noncoherent detectors. AM is easily demodulated
using an envelope detector whereas FM is demodulated using a discriminator or
slope detector. In FM the instantaneous frequency of the carrier signal is varied
linearly with the baseband message signal m(t), as shown in following equation:
sFM(t) = Ac cos[2 fct + (t)] = Ac cos[2 fct + 2 kf
Z
m( )d ] (6.30)
where Ac, is the amplitude of the carrier, fc is the carrier frequency, and kf is the
frequency deviation constant (measured in units of Hz/V).
Phase modulation (PM) is a form of angle modulation in which the angle (t) of
the carrier signal is varied linearly with the baseband message signal m(t), as shown
in equation below.
sPM(t) = Ac cos(2 fct + k m(t)) (6.31)
The frequency modulation index f , de nes the relationship between the message
amplitude and the bandwidth of the transmitted signal, and is given by
f = kfAm
W
=

W
(6.32)
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where Am is the peak value of the modulating signal, f is the peak frequency
deviation of the transmitter and W is the maximum bandwidth of the modulating
signal.
The phase modulation index p is given by
p = k Am = (6.33)
where, is the peak phase deviation of the transmitter.
6.8.2 BFSK
In Binary Frequency Shift keying (BFSK),the frequency of constant amplitude carrier
signal is switched between two values according to the two possible message
states (called high and low tones) corresponding to a binary 1 or 0. Depending on
how the frequency variations are imparted into the transmitted waveform,the FSK
signal will have either a discontinuous phase or continuous phase between bits. In
general, an FSK signal may be represented as
S(t) =
q
(2Eb=T ) cos(2 fit): (6.34)
where T is the symbol duration and Eb is the energy per bit.
Si =
q
(Eb) (t): (6.35)
(t) =
q
(2=T ) cos(2 fit): (6.36)
There are two FSK signals to represent 1 and 0, i.e.,
S1(t) =
q
(2Eb=T ) cos(2 f1t + (0)) ! 1 (6.37)
S2(t) =
q
(2Eb=T ) cos(2 f2t + (0)) ! 0 (6.38)
where (0) sums the phase up to t = 0. Let us now consider a continuous phase
FSK as
S(t) =
q
(2Eb=T ) cos(2 fct + (t)): (6.39)
Expressing (t) in terms of (0) with a new unknown factor h, we get
(t) = (0) ht=T 0 t T (6.40)
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and therefore
S(t) =
s
2Eb
T
cos(2 fct ht=T + (0)) =
s
2Eb
T
cos(2 (fc h=2T)t + (0)):(6.41)
It shows that we can choose two frequencies f1 and f2 such that
f1 = fc + h=2T (6.42)
f2 = fc 􀀀 h=2T (6.43)
for which the expression of FSK conforms to that of CPFSK. On the other hand, fc
and h can be expressed in terms of f1 and f2 as
fc = [f1 + f2]=2 (6.44)
h =
(f1 􀀀 f2)
1=T
: (6.45)
Therefore, the unknown factor h can be treated as the di erence between f1 and f2,
normalized with respect to bit rate 1=T . It is called the deviation ratio. We know
that (t) 􀀀 (0) = ht=T , 0 t T. If we substitute t = T, we have
(T) 􀀀 (0) = h where (6.46)
= h ! 1 (6.47)
= 􀀀 h ! 0 (6.48)
This type of CPFSK is advantageous since by looking only at the phase, the transmitted
bit can be predicted. In Figure 6.8, we show a phase tree of such a CPFSK
signal with the transmitted bit stream of 1101000.
A special case of CPFSK is achieved with h = 0:5, and the resulting scheme is
called Minimum Shift Keying (MSK) which is used in mobile communications. In
this case, the phase di erences reduce to only =2 and the phase tree is called the
phase trellis. An MSK signal can also be thought as a special case of OQPSK where
the baseband rectangular pulses are replaced by half sinusoidal pulses. Spectral
characteristics of an MSK signal is shown in Figure 6.9 from which it is clear that
ACI is present in the spectrum. Hence a pulse shaping technique is required. In
order to have a compact signal spectrum as well as maintaining the constant envelope
property, we use a pulse shaping lter with
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Figure 6.8: Phase tree of 1101000 CPFSK sequence.
Figure 6.9: Spectrum of MSK
1. a narrow BW frequency and sharp cuto characteristics (in order to suppress
the high frequency component of the signal);
2. an impulse response with relatively low overshoot (to limit FM instant frequency
deviation;
3. a phase trellis with =2 for odd T and 0 or values for even T.
6.9 GMSK Scheme
GMSK is a simple modulation scheme that may be taken as a derivative of MSK.
In GMSK, the sidelobe levels of the spectrum are further reduced by passing a non-
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Figure 6.10: GMSK generation scheme.
return to zero (NRZ-L) data waveform through a premodulation Gaussian pulse
shaping lter. Baseband Gaussian pulse shaping smoothes the trajectory of the
MSK signals and hence stabilizes instantaneous frequency variations over time. This
has the e ect of considerably reducing the sidelobes in the transmitted spectrum.
A GMSK generation scheme with NRZ-L data is shown in Figure 6.10 and a receiver
of the same scheme with some MSI gates is shown in Figure 6.11.
6.10 GMSK Generator
The GMSK premodulation lter has characteristic equation given by
H(f) = exp(􀀀(ln 2=2)(f=B)2) (6.49)
H(f) = exp(􀀀( f)2)
where,
( )2 = ln 2=2(1=B)2: (6.50)
The premodulation Gaussian ltering introduces ISI in the transmitted signal, but
it can be shown that the degradation is not that great if the 3dB bandwidth-bit
duration product (BT) is greater than 0.5.
Spectrum of GMSK scheme is shown in Figure 6.12. From this gure, it is evident
that when we are decreasing BT product, the out of band response decreases but
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Figure 6.11: A simple GMSK receiver.
on the other hand irreducible error rate of the LPF for ISI increases. Therefore, a
compromise between these two is required.
Problem: Find the 3dB BW for a Gaussian LPF used to produce 0.25 GMSK
with a channel data rate Rb=270 kbps.What is the 90 percent power BW of the RF
lter?
Solution: From the problem statement it is clear that
T = 1=Rb = 1=270 (103) = 3:7 sec (6.51)
Solving for B where BT = 0:25,
B = 0:25=T = 67:567kHz (6.52)
Thus the 3 - dB bandwidth is 67.567 kHz. We use below table g 6 to nd out that
90 % power bandwidth is 0.57 Rb.
90 % RF BW = 0:57Rb = 153:9 kHz.
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Figure 6.12: Spectrum of GMSK scheme.
6.11 Two Practical Issues of Concern
6.11.1 Inter Channel Interference
In FDMA, subscribers are allotted frequency slots called channels in a given band
of the electromagnetic spectrum. The side lobes generated due to the transmission
of a symbol in a particular channel overlaps with the channels placed adjacently.
This is because of the fact that transmission of a time limited pulse leads to spectral
spreading in the frequency domain. During simultaneous use of adjacent channels,
when there is signi cant amount of power present in the side lobes, this kind of
interference becomes so severe that the required symbol in a particular frequency
slot is completely lost.
Moreover if two terminals transmit equal power then due to wave propagation
through di erent distances to the receiver, the received signal levels in the two
frequency slots will di er greatly. In such a case the side lobes of the stronger signal
will severely degrade the transmitted signal in the next frequency slot having low
power level. This is known as the near far problem.
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6.11.2 Power Ampli er Nonlinearity
Power ampli ers may be designed as class A, class B, class AB, class C and class D.
They form an essential section of mobile radio terminals. Due to power constraints
on a transmitting terminal, an e cient power ampli er is required which can convert
most of the input power to RF power. Class A ampli er is a linear ampli er but
it has a power e ciency of only 25 %. As we go for subsequent ampli ers having
greater power e ciency, the nonlinearity of the ampli er increases.
In general, an ampli er has linear input output characteristics over a range
of input signal level, that is, it has a constant gain. However, beyond an input
threshold level, the gain of the ampli er starts decreasing. Thus the amplitude of
a signal applied at the input of an ampli er su ers from amplitude distortion and
the resulting waveform obtained at the output of the ampli er is of the form of
an amplitude modulated signal. Similarly, the phase characteristic of a practical
ampli er is not constant over all input levels and results in phase distortion of the
form of phase modulation.
The operating point of a practical ampli er is given in terms of either the input
back-o or the output back-o .
Input back 􀀀 off = 10 log1 0

Vin;rms
Vout;rms
!
(6.53)
Output back 􀀀 off = 10 log1 0

Vout;rms
Vout;rms
!
(6.54)
6.12 Receiver performance in multipath channels
For a
at fading channel, the probability of error for coherent BPSK and coherent
BFSK are respectively given as,
Pe;BPSK =
1
2

1 􀀀
r


1 +


(6.55)
Pe;BFSK =
1
2

1 􀀀
r


2 +


(6.56)
(6.57)
where
 is given by,

 = Eb
N0
E( 2) (6.58)
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2 represents the instantaneous power values of the Rayleigh fading channel and E
denotes the expectation operator.
Similarly, for di erential BPSK and non coherent BFSK probability of error
expressions are
Pe;DPSK =
1
2(1 +
)
(6.59)
Pe;NCFSK =
1
(2 +
) : (6.60)
For large values of SNR = Eb
N0
the error probability given above have the simpli ed
expression.
Pe;BPSK =
1
4

(6.61)
Pe;BFSK =
1
2

(6.62)
Pe;DPSK =
1
2

(6.63)
Pe;NCFSK =
1


: (6.64)
From the above equations we observe that an inverse algebraic relation exists between
the BER and SNR. This implies that if the required BER range is around
10􀀀3 to 10􀀀6, then the SNR range must be around 30dB to 60dB.
6.12.1 Bit Error Rate and Symbol Error Rate
Bit error rate (Peb) is the same as symbol error rate (Pes) when a symbol consists
of a single bit as in BPSK modulation. For an MPSK scheme employing gray coded
modulation, where N bits are mapped to a one of the M symbols, such that 2N = M,
Peb is given by
Peb
Pes
log2M
(6.65)
And for M-ary orthogonal signalling Peb is given by
Peb = M=2
M 􀀀 1Pes: (6.66)
6.13 Example of a Multicarrier Modulation: OFDM
Multiplexing is an important signal processing operation in which a number of signals
are combined and transmitted parallelly over a common channel. In order to
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avoid interference during parallel transmission, the signals can be separated in frequency
and then the resulting technique is called Frequency Division Multiplexing
(FDM). In FDM, the adjacent bands are non overlapping but if overlap is allowed by
transmitting signals that are mutually orthogonal (that is, there is a precise mathematical
relationship between the frequencies of the transmitted signals) such that
one signal has zero e ect on another, then the resulting transmission technique is
known as Orthogonal Frequency Division Multiplexing (OFDM).
OFDM is a technique of transmitting high bit rate data into several parallel
streams of low bit rate data. At any instant, the data transmitted simultaneously
in each of these parallel data streams is frequency modulated by carriers (called
subcarriers) which are orthogonal to each other. For high data rate communication
the bandwidth (which is limited) requirement goes on increasing as the data rate
increases or the symbol duration decreases. Thus in OFDM, instead of sending a
particular number of symbols, say P, in T seconds serially, the P symbols can be
sent in parallel with symbol duration now increased to T seconds instead of T/P
seconds as was previously.
This o ers many advantages in digital data transmission through a wireless time
varying channel. The primary advantage of increasing the symbol duration is that
the channel experiences
at fading instead of frequency selective fading since it is
ensured that in the time domain the symbol duration is greater than the r.m.s.
delay spread of the channel. Viewed in the frequency domain this implies that the
bandwidth of the OFDM signal is less than coherent bandwidth of the channel.
Although the use of OFDM was initially limited to military applications due to
cost and complexity considerations, with the recent advances in large-scale highspeed
DSP, this is no longer a major problem. This technique is being used, in
digital audio broadcasting (DAB), high de nition digital television broadcasting
(HDTV), digital video broadcasting terrestrial TV (DVB-T), WLAN systems based
on IEEE 802.11(a) or HiperLan2, asymmetric digital subscriber lines (ADSL) and
mobile communications. Very recently, the signi cance of the COFDM technique for
UWA (underwater acoustic channel) has also been indicated. Moreover related or
combined technology such as CDMA-OFDM, TDMA-OFDM, MIMO-OFDM, Vector
OFDM (V-OFDM), wide-band OFDM (W-OFDM),
ash OFDM (F-OFDM),
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OFDMA, wavelet-OFDM have presented their great advantages in certain application
areas.
6.13.1 Orthogonality of Signals
Orthogonal signals can be viewed in the same perspective as we view vectors which
are perpendicular/orthogonal to each other. The inner product of two mutually
orthogonal vectors is equal to zero. Similarly the inner product of two orthogonal
signals is also equal to zero.
Let  k(t) = ej2 fkt and  n(t) = ej2 fnt be two complex exponential signals whose
inner product, over the time duration of Ts, is given by:
N =
Z (i+1)Ts
iTs
 k(t):
n(t)dt (6.67)
When this integral is evaluated, it is found that if fk and fn are integer multiples
of 1/Ts then N equals zero. This implies that for two harmonics of an exponential
function having a fundamental frequency of 1/Ts, the inner product becomes zero
.But if fk = fn then N equals Ts which is nothing but the energy of the complex
exponential signal in the time duration of Ts.
6.13.2 Mathematical Description of OFDM
Let us now consider the simultaneous or parallel transmission of P number of complex
symbols in the time slot of Ts second (OFDM symbol time duration) and a set
of P orthogonal subcarriers, such that each subcarrier gets amplitude modulated
by a particular symbol from this set of P symbols. Let each orthogonal carrier
be of the form exp

j2 n t
Ts

, where n varies as 0; 1; 2::(P 􀀀 1). Here the variable
`n' denotes the nth parallel path corresponding to the nth subcarrier. Mathematically,
we can obtain the transmitted signal in Ts seconds by summing up all the P
number of amplitude modulated subcarriers, thereby yielding the following equation:
p(t) =
PX􀀀1
n=0
cngn(t)exp

j2 n
t
Ts

for 0 t Ts (6.68)
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If p(t) is sampled at t = kTs=P, then the resulting waveform, is:
p(k) =
PX􀀀1
n=0
cngn(kTs=P)exp

j2 n
kTs=P
Ts

=
1
p
Ts
PX􀀀1
n=0
cnexp

j2 n
k
P

for 0 k P 􀀀 1 (6.69)
This is nothing but the IDFT on the symbol block of P symbols. This can be realized
using IFFT but the constraint is that P has to be a power of 2. So at the receiver,
FFT can be done to get back the required block of symbols. This implementation is
better than using multiple oscillators for subcarrier generation which is uneconomical
and since digital technology has greatly advanced over the past few decades, IFFTs
and FFTs can be implemented easily. The frequency spectrum, therefore consists
of a set of P partially overlapping sinc pulses during any time slot of duration Ts.
This is due to the fact that the Fourier Transform of a rectangular pulse is a sinc
function. The receiver can be visualized as consisting of a bank of demodulators,
translating each subcarrier down to DC, then integrating the resulting signal over a
symbol period to recover the raw data.
But the OFDM symbol structure so generated at the transmitter end needs to
be modi ed. Since inter symbol interference (ISI) is introduced by the transmission
channel due to multipaths and also due to the fact that when the bandwidth of
OFDM signal is truncated, its e ect in the time domain is to cause symbol spreading
such that a part of the symbol overlaps with the adjacent symbols. In order to cope
with ISI as discussed previously the OFDM symbol duration can be increased. But
this might not be feasible from the implementation point of view speci cally in terms
of FFT size and Doppler shifts.
A di erent approach is to keep a guard time interval between two OFDM symbols
in which part of the symbol is copied from the end of the symbol to the front and is
popularly known as the cyclic-pre x. If we denote the guard time interval as Tg and
Ts be the useful symbol duration, then after this cyclical extension the total symbol
duration becomes T = Tg + Ts. When the guard interval is longer than the length
of the channel impulse response, or the multipath delay, then ISI can be eliminated.
However the disadvantage is the reduction in data rate or throughput and greater
power requirements at the transmitting end. The OFDM transmitter and receiver
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Figure 6.13: OFDM Transmitter and Receiver Block Diagram.
sections are as given in the following diagram.
6.14 Conclusion
In this chapter, a major chunk has been devoted to digital communication systems
which obviously have certain distinction in comparison to their analog counterpart
due to their signal-space representation. The important modulation techniques for
wireless communication such as QPSK, MSK, GMSK were taken up at length. A
relatively new modulation technology, OFDM, has also been discussed. Certain
practical issues of concern are also discussed. It should be noted that albeit implementing
these e cient modulation techniques, the channel still introduces fading in
di erent ways. In order to prevent that, we need some additional signal processing
techniques mainly at the receiver side. These techniques are discussed in the next
chapter.
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6.15 References
1. B. P. Lathi and Z. Ding, Modern Digital and Analog Communication Systems,
4th ed. NY: Oxford University Press, 2009.
2. B. Sklar, Digital Communications: Fundamentals and Applications, 2nd ed.
Singapore: Pearson Education, Inc., 2005.
3. R. Blake, Electronic Communication Systems. Delmar, Singapore: Thomson
Asia Pvt Ltd, 2002.
4. J. G. Proakis and M. Salehi, Communication Systems Engineering, 2nd ed.
Singapore: Pearson Education, Inc., 2002.
5. T. S. Rappaport, Wireless Communications: Principles and Practice, 2nd ed.
Singapore: Pearson Education, Inc., 2002.
6. S. Haykin and M. Moher, Modern Wireless Communications. Singapore: Pearson
Education, Inc., 2002.
7. W. H. Tranter et. al., Principles of Communication Systems Simulation. Singapore:
Pearson Education, Inc., 2004.
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Chapter 7
Techniques to Mitigate Fading
E ects
7.1 Introduction
Apart from the better transmitter and receiver technology, mobile communications
require signal processing techniques that improve the link performance. Equalization,
Diversity and channel coding are channel impairment improvement techniques.
Equalization compensates for Inter Symbol Interference (ISI) created by multipath
within time dispersive channels. An equalizer within a receiver compensates for
the average range of expected channel amplitude and delay characteristics. In other
words, an equalizer is a lter at the mobile receiver whose impulse response is inverse
of the channel impulse response. As such equalizers nd their use in frequency selective
fading channels. Diversity is another technique used to compensate fast fading
and is usually implemented using two or more receiving antennas. It is usually employed
to reduce the depths and duration of the fades experienced by a receiver in
a
at fading channel. Channel coding improves mobile communication link performance
by adding redundant data bits in the transmitted message.At the baseband
portion of the transmitter, a channel coder maps a digital message sequence in to
another speci c code sequence containing greater number of bits than original contained
in the message. Channel Coding is used to correct deep fading or spectral
null. We discuss all three of these techniques in this chapter. A general framework
of the fading e ects and their mitigation techniques is shown in Figure 7.1.
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Figure 7.1: A general framework of fading e ects and their mitigation techniques.
7.2 Equalization
ISI has been identi ed as one of the major obstacles to high speed data transmission
over mobile radio channels. If the modulation bandwidth exceeds the coherence
bandwidth of the radio channel (i.e., frequency selective fading), modulation pulses
are spread in time, causing ISI. An equalizer at the front end of a receiver compensates
for the average range of expected channel amplitude and delay characteristics.
As the mobile fading channels are random and time varying, equalizers must track
the time-varying characteristics of the mobile channel and therefore should be timevarying
or adaptive. An adaptive equalizer has two phases of operation: training
and tracking. These are as follows.
Training Mode:
Initially a known, xed length training sequence is sent by the transmitter so
that the receiver equalizer may average to a proper setting.
Training sequence is typically a pseudo-random binary signal or a xed, of
prescribed bit pattern.
The training sequence is designed to permit an equalizer at the receiver to
acquire the proper lter coe cient in the worst possible channel condition.
An adaptive lter at the receiver thus uses a recursive algorithm to evaluate
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the channel and estimate lter coe cients to compensate for the channel.
Tracking Mode:
When the training sequence is nished the lter coe cients are near optimal.
Immediately following the training sequence, user data is sent.
When the data of the users are received, the adaptive algorithms of the equalizer
tracks the changing channel.
As a result, the adaptive equalizer continuously changes the lter characteristics
over time.
7.2.1 A Mathematical Framework
The signal received by the equalizer is given by
x(t) = d(t) h (t) + nb (t) (7.1)
where d(t) is the transmitted signal, h(t) is the combined impulse response of the
transmitter,channel and the RF/IF section of the receiver and nb (t) denotes the
baseband noise.
If the impulse response of the equalizer is heq (t), the output of the equalizer is
^y (t) = d (t) h (t) heq (t) + nb (t) heq (t) = d (t) g (t) + nb (t) heq (t) : (7.2)
However, the desired output of the equalizer is d(t) which is the original source data.
Assuming nb (t)=0, we can write y(t) = d(t), which in turn stems the following
equation:
g (t) = h (t) heq (t) = (t) (7.3)
The main goal of any equalization process is to satisfy this equation optimally. In
frequency domain it can be written as
Heq (f)H (f) = 1 (7.4)
which indicates that an equalizer is actually an inverse lter of the channel. If the
channel is frequency selective, the equalizer enhances the frequency components with
small amplitudes and attenuates the strong frequencies in the received frequency
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spectrum in order to provide a
at, composite received frequency response and
linear phase response. For a time varying channel, the equalizer is designed to track
the channel variations so that the above equation is approximately satis ed.
7.2.2 Zero Forcing Equalization
In a zero forcing equalizer, the equalizer coe cients cn are chosen to force the samples
of the combined channel and equalizer impulse response to zero. When each of the
delay elements provide a time delay equal to the symbol duration T, the frequency
response Heq (f) of the equalizer is periodic with a period equal to the symbol rate
1/T. The combined response of the channel with the equalizer must satisfy Nyquist's
criterion
Hch (f)Heq (f) = 1; jfj < 1/2T (7.5)
where Hch (f) is the folded frequency response of the channel. Thus, an in nite
length zero-forcing ISI equalizer is simply an inverse lter which inverts the folded
frequency response of the channel.
Disadvantage: Since Heq (f) is inverse of Hch (f) so inverse lter may excessively
amplify the noise at frequencies where the folded channel spectrum has high attenuation,
so it is rarely used for wireless link except for static channels with high SNR
such as local wired telephone. The usual equalizer model follows a time varying or
adaptive structure which is given next.
7.2.3 A Generic Adaptive Equalizer
The basic structure of an adaptive lter is shown in Figure 7.2. This lter is called
the transversal lter, and in this case has N delay elements, N+1 taps and N+1
tunable complex multipliers, called weights. These weights are updated continuously
by an adaptive algorithm. In the gure the subscript k represents discrete time
index. The adaptive algorithm is controlled by the error signal ek. The error signal
is derived by comparing the output of the equalizer, with some signal dk which is
replica of transmitted signal. The adaptive algorithm uses ek to minimize the cost
function and uses the equalizer weights in such a manner that it minimizes the cost
function iteratively. Let us denote the received sequence vector at the receiver and
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Figure 7.2: A generic adaptive equalizer.
the input to the equalizer as
xk = [xk; xk􀀀1; :::::; xk􀀀N]T ; (7.6)
and the tap coe cient vector as
wk = [w0
k;w1
k; :::::;wN
k ]T : (7.7)
Now, the output sequence of the equalizer yk is the inner product of xk and wk, i.e.,
yk = hxk;wki = xT
k wk = wT
k xk: (7.8)
The error signal is de ned as
ek = dk 􀀀 yk = dk 􀀀 xT
k wk: (7.9)
Assuming dk and xk to be jointly stationary, the Mean Square Error (MSE) is given
as
MSE = E[e2
k] = E[(dk 􀀀 yk)2]
= E[(dk 􀀀 xT
k wk)2]
= E[d2
k] + wT
k E[xkxT
k ]wk 􀀀 2E[dkxT
k ]wk (7.10)
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where wk is assumed to be an array of optimum values and therefore it has been
taken out of the E() operator. The MSE then can be expressed as
MSE = = 2
k + wT
k Rwk 􀀀 2pTwk (7.11)
where the signal variance 2
d = E[d2
k] and the cross correlation vector p between the
desired response and the input signal is de ned as
p = E [dkxk] = E

dkxk dkxk􀀀1 dkxk􀀀2 dkxk􀀀N

: (7.12)
The input correlation matrix R is de ned as an (N + 1) (N + 1) square matrix,
where
R = E
h
xkxT
k
i
= E
2
666666666664
x2
k xkxk􀀀1 xkxk􀀀2 xkxk􀀀N
xk􀀀1xk x2
k􀀀1 xk􀀀1xk􀀀2 xk􀀀1xk􀀀N
xk􀀀2xk xk􀀀2xk􀀀1 x2
k􀀀2 xk􀀀2xk􀀀N
...
...
...

...
xk􀀀Nxk xk􀀀Nxk􀀀1 xk􀀀Nxk􀀀2 x2
k􀀀N
3
777777777775
: (7.13)
Clearly, MSE is a function of wk. On equating @
@wk
to 0, we get the condition for
minimum MSE (MMSE) which is known as Wiener solution:
wk = R􀀀1p: (7.14)
Hence, MMSE is given by the equation
MMSE = min = 2
d 􀀀 pTwk: (7.15)
7.2.4 Choice of Algorithms for Adaptive Equalization
Since an adaptive equalizer compensates for an unknown and time varying channel,
it requires a speci c algorithm to update the equalizer coe cients and track the
channel variations. Factors which determine algorithm's performance are:
Rate of convergence: Number of iterations required for an algorithm, in response
to a stationary inputs, to converge close enough to optimal solution. A fast
rate of convergence allows the algorithm to adapt rapidly to a stationary environment
of unknown statistics.
Misadjustment: Provides a quantitative measure of the amount by which the
nal value of mean square error, averaged over an ensemble of adaptive lters,
deviates from an optimal mean square error.
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Computational complexity: Number of operations required to make one complete
iteration of the algorithm.
Numerical properties: Inaccuracies like round-o noise and representation
errors in the computer, which in
uence the stability of the algorithm.
Three classic equalizer algorithms are primitive for most of today's wireless standards.
These include the Zero Forcing Algorithm (ZF), the Least Mean Square Algorithm
(LMS), and the Recursive Least Square Algorithm (RLS). Below, we discuss
a few of the adaptive algorithms.
Least Mean Square (LMS) Algorithm
LMS algorithm is the simplest algorithm based on minimization of the MSE between
the desired equalizer output and the actual equalizer output, as discussed earlier.
Here the system error, the MSE and the optimal Wiener solution remain the same
as given the adaptive equalization framework.
In practice, the minimization of the MSE is carried out recursively, and may be
performed by use of the stochastic gradient algorithm. It is the simplest equalization
algorithm and requires only 2N+1 operations per iteration. The lter weights are
updated by the update equation. Letting the variable n denote the sequence of
iteration, LMS is computed iteratively by
wk (n + 1) = wk (n) + ek (n) x (n 􀀀 k) (7.16)
where the subscript k denotes the kth delay stage in the equalizer and is the step
size which controls the convergence rate and stability of the algorithm.
The LMS equalizer maximizes the signal to distortion ratio at its output within
the constraints of the equalizer lter length. If an input signal has a time dispersion
characteristics that is greater than the propagation delay through the equalizer, then
the equalizer will be unable to reduce distortion. The convergence rate of the LMS
algorithm is slow due to the fact that there is only one parameter, the step size, that
controls the adaptation rate. To prevent the adaptation from becoming unstable,
the value of is chosen from
0 < < 2
,
XN
i=1
i (7.17)
where i is the i-th eigenvalue of the covariance matrix R.
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Normalized LMS (NLMS) Algorithm
In the LMS algorithm, the correction that is applied to wk (n) is proportional to
the input sample x (n 􀀀 k). Therefore when x (n 􀀀 k) is large, the LMS algorithm
experiences gradient noise ampli cation. With the normalization of the LMS step
size by kx (n)k2 in the NLMS algorithm, this problem is eliminated. Only when
x(n􀀀k) becomes close to zero, the denominator term kx (n)k2 in the NLMS equation
becomes very small and the correction factor may diverge. So, a small positive
number " is added to the denominator term of the correction factor. Here, the step
size is time varying and is expressed as
(n) =
kx (n)k2 + "
: (7.18)
Therefore, the NLMS algorithm update equation takes the form of
wk (n + 1) = wk (n) +
kx (n)k2 + "
ek (n) x (n 􀀀 k) : (7.19)
7.3 Diversity
Diversity is a method used to develop information from several signals transmitted
over independent fading paths. It exploits the random nature of radio propagation
by nding independent signal paths for communication. It is a very simple concept
where if one path undergoes a deep fade, another independent path may have a
strong signal. As there is more than one path to select from, both the instantaneous
and average SNRs at the receiver may be improved. Usually diversity decisions are
made by receiver. Unlike equalization, diversity requires no training overhead as a
training sequence is not required by transmitter. Note that if the distance between
two receivers is a multiple of =2, there might occur a destructive interference between
the two signals. Hence receivers in diversity technique are used in such a
way that the signal received by one is independent of the other. Diversity can be of
various forms, starting from space diversity to time diversity. We take up the types
one by one in the sequel.
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Figure 7.3: Receiver selection diversity, with M receivers.
7.3.1 Di erent Types of Diversity
Space Diversity
A method of transmission or reception, or both, in which the e ects of fading are
minimized by the simultaneous use of two or more physically separated antennas,
ideally separated by one half or more wavelengths. Signals received from spatially
separated antennas have uncorrelated envelopes.
Space diversity reception methods can be classi ed into four categories: selection,
feedback or scanning, maximal ratio combining and equal gain combining.
(a) Selection Diversity:
The basic principle of this type of diversity is selecting the best signal among all
the signals received from di erent branches at the receiving end. Selection Diversity
is the simplest diversity technique. Figure 7.3 shows a block diagram of this method
where 'M' demodulators are used to provide M diversity branches whose gains are
adjusted to provide the same average SNR for each branch. The receiver branches
having the highest instantaneous SNR is connected to the demodulator.
Let M independent Rayleigh fading channels are available at a receiver. Each
channel is called a diversity branch and let each branch has the same average SNR.
The signal to noise ratio is de ned as
SNR = 􀀀 = Eb
N0
2 (7.20)
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where Eb is the average carrier energy, N0 is the noise PSD, is a random variable
used to represent amplitude values of the fading channel.
The instantaneous SNR(
i) is usually de ned as
i = instantaneous signal power
per branch/mean noise power per branch. For Rayleigh fading channels, has a
Rayleigh distribution and so 2 and consequently
i have a chi-square distribution
with two degrees of freedom. The probability density function for such a channel is
p (
i) =
1
􀀀e
􀀀
i
􀀀 : (7.21)
The probability that any single branch has an instantaneous SNR less than some
de ned threshold
 is
Pr [
i
] =
Z

0
p (
i) d
i =
Z

0
1
􀀀e
􀀀
i
􀀀 d
i = 1 􀀀 e
􀀀

􀀀 = P(􀀀): (7.22)
Similarly, the probability that all M independent diversity branches receive signals
which are simultaneously less than some speci c SNR threshold
 is
Pr [
1;
2; : : : ;
M
] =

1 􀀀 e
􀀀

􀀀
M
= PM (
) (7.23)
where PM (
) is the probability of all branches failing to achieve an instantaneous
SNR =
. Quite clearly, PM(􀀀) < P(􀀀). If a single branch achieves SNR >
, then
the probability that SNR >
 for one or more branches is given by
Pr [
i >
] = 1 􀀀 PM (
) = 1 􀀀

1 􀀀 e
􀀀

􀀀
M
(7.24)
which is more than the required SNR for a single branch receiver. This expression
shows the advantage when a selection diversity is used.
To determine of average signal to noise ratio, we rst nd out the pdf of
 as
pM (
) = d
d

PM (
) = M
􀀀

1 􀀀 e􀀀
/􀀀
M􀀀1
e􀀀
/􀀀: (7.25)
The average SNR,
, can be then expressed as

 =
1 Z
0

pM (
) d
 = 􀀀
1 Z
0
Mx
􀀀
1 􀀀 e􀀀x
M􀀀1
e􀀀xdx (7.26)
where x =
/􀀀 and 􀀀 is the average SNR for a single branch, when no diversity is
used.
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This equation shows an average improvement in the link margin without requiring
extra transmitter power or complex circuitry, and it is easy to implement as
it needed a monitoring station and an antenna switch at the receiver. It is not an
optimal diversity technique as it doesn't use all the possible branches simultaneously.
(b) Feedback or Scanning Diversity:
Scanning all the signals in a xed sequence until the one with SNR more than a
predetermined threshold is identi ed. Feedback or scanning diversity is very similar
to selection diversity except that instead of always using the best of N signals, the N
signals are scanned in a xed sequence until one is found to be above a predetermined
threshold. This signal is then received until it falls below threshold and the scanning
process is again initiated. The resulting fading statistics are somewhat inferior, but
the advantage is that it is very simple to implement(only one receiver is required).
(c) Maximal Ratio Combining:
Signals from all of the m branches are weighted according to their individual
signal voltage to noise power ratios and then summed. Individual signals must be
cophased before being summed, which generally requires an individual receiver and
phasing circuit for each antenna element. Produces an output SNR equal to the
sum of all individual SNR. Advantage of producing an output with an acceptable
SNR even when none of the individual signals are themselves acceptable. Modern
DSP techniques and digital receivers are now making this optimal form, as it gives
the best statistical reduction of fading of any known linear diversity combiner. In
terms of voltage signal,
rm =
mX
i=1
Giri (7.27)
where Gi is the gain and ri is the voltage signal from each branch.
(d) Equal Gain Combining:
In some cases it is not convenient to provide for the variable weighting capability
required for true maximal ratio combining. In such cases, the branch weights are
all set unity, but the signals from each branch are co-phased to provide equal gain
combining diversity. It allows the receiver to exploit signals that are simultaneously
received on each branch. Performance of this method is marginally inferior to maximal
ratio combining and superior to Selection diversity. Assuming all the Gi to be
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Figure 7.4: Maximal ratio combining technique.
unity, here,
rm =
mX
i=1
ri: (7.28)
Polarization Diversity
Polarization Diversity relies on the decorrelation of the two receive ports to achieve
diversity gain. The two receiver ports must remain cross-polarized. Polarization
Diversity at a base station does not require antenna spacing. Polarization diversity
combines pairs of antennas with orthogonal polarizations (i.e. horizontal/vertical,
slant 45o, Left-hand/Right-hand CP etc). Re
ected signals can undergo polarization
changes depending on the channel. Pairing two complementary polarizations, this
scheme can immunize a system from polarization mismatches that would otherwise
cause signal fade. Polarization diversity has prove valuable at radio and mobile com-
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munication base stations since it is less susceptible to the near random orientations
of transmitting antennas.
Frequency Diversity
In Frequency Diversity, the same information signal is transmitted and received
simultaneously on two or more independent fading carrier frequencies. Rationale
behind this technique is that frequencies separated by more than the coherence
bandwidth of the channel will be uncorrelated and will thus not experience the same
fades. The probability of simultaneous fading will be the product of the individual
fading probabilities. This method is employed in microwave LoS links which carry
several channels in a frequency division multiplex mode (FDM). Main disadvantage
is that it requires spare bandwidth also as many receivers as there are channels used
for the frequency diversity.
Time Diversity
In time diversity, the signal representing the same information are sent over the
same channel at di erent times. Time diversity repeatedly transmits information at
time spacings that exceeds the coherence time of the channel. Multiple repetition
of the signal will be received with independent fading conditions, thereby providing
for diversity. A modern implementation of time diversity involves the use of RAKE
receiver for spread spectrum CDMA, where the multipath channel provides redundancy
in the transmitted message. Disadvantage is that it requires spare bandwidth
also as many receivers as there are channels used for the frequency diversity. Two
important types of time diversity application is discussed below.
Application 1: RAKE Receiver
In CDMA spread spectrum systems, CDMA spreading codes are designed to provide
very low correlation between successive chips, propagation delay spread in the radio
channel provides multiple version of the transmitted signal at the receiver. Delaying
multipath components by more than a chip duration, will appear like uncorrelated
noise at a CDMA receiver. CDMA receiver may combine the time delayed versions
of the original signal to improve the signal to noise ratio at the receiver. RAKE
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Figure 7.5: RAKE receiver.
receiver collect the time shifted versions of the original signal by providing a separate
correlation receiver for M strongest multipath components. Outputs of each
correlator are weighted to provide a better estimate of the transmitted signal than
provided by a single component. Demodulation and bit decisions are based on the
weighted output of the correlators. Schematic of a RAKE receiver is shown in Figure
7.5.
Application 2: Interleaver
In the encoded data bits, some source bits are more important than others, and
must be protected from errors. Many speech coder produce several important bits
in succession. Interleaver spread these bit out in time so that if there is a deep fade
or noise burst, the important bits from a block of source data are not corrupted
at the same time. Spreading source bits over time, it becomes possible to make
use of error control coding. Interleaver can be of two forms, a block structure or a
convolutional structure.
A block interleaver formats the encoded data into a rectangular array of m rows
and n columns, and interleaves nm bits at a time. Each row contains a word of
source data having n bits. an interleaver of degree m consists of m rows. source bits
are placed into the interleaver by sequentially increasing the row number for each
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successive bit, and forming the columns. The interleaved source data is then read
out row-wise and transmitted over the channel. This has the e ect of separating
the original source bits by m bit periods. At the receiver, de-interleaver stores the
received data by sequentially increasing the row number of each successive bit, and
then clocks out the data row-wise, one word at a time. Convolutional interleavers
are ideally suited for use with convolutional codes.
7.4 Channel Coding
In channel coding, redundant data bits are added in the transmitted message so
that if an instantaneous fade occurs in the channel, the data may still be recovered
at the receiver without the request of retransmission. A channel coder maps
the transmitted message into another speci c code sequence containing more bits.
Coded message is then modulated for transmission in the wireless channel. Channel
Coding is used by the receiver to detect or correct errors introduced by the channel.
Codes that used to detect errors, are error detection codes. Error correction codes
can detect and correct errors.
7.4.1 Shannon's Channel Capacity Theorem
In 1948, Shannon showed that by proper encoding of the information, errors induced
by a noise channel can be reduced to any desired level without sacri cing the rate
of information transfer. Shannon's channel capacity formula is applicable to the
AWGN channel and is given by:
C = B log2

1 + S
N

= B log2

1 + P
N0B

= B log2

1 + EbRb
N0B

(7.29)
where C is the channel capacity (bit/s), B is the channel bandwidth (Hz), P is the
received signal power (W), N0 is the single sided noise power density (W/Hz), Eb is
the average bit energy and Rb is transmission bit rate.
Equation (7.29) can be normalized by the bandwidth B and is given as
C
B
= log2

1 + EbRb
N0B

(7.30)
and the ratio C/B is denoted as bandwidth e ciency. Introduction of redundant
bits increases the transmission bit rate and hence it increases the bandwidth requirement,
which reduces the bandwidth e ciency of the link in high SNR conditions, but
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provides excellent BER performance at low SNR values. This leads to the following
two inferences.
Corollary 1 : While dealing within maximum channel capacity, introduction of redundant
bits increase the transmitter rate and hence bandwidth requirement also
increases, while decreasing the bandwidth e ciency, but it also decreases the BER.
Corollary 2 : If data redundancy is not introduced in a wideband noisy environment,
error free performance in not possible (for example, CDMA communication in 3G
mobile phones).
A channel coder operates on digital message (or source) data by encoding the source
information into a code sequence for transmission through the channel. The error
correction and detection codes are classi ed into three groups based on their structure.
1. Block Code
2. Convolution Code
3. Concatenated Code.
7.4.2 Block Codes
Block codes are forward error correction (FEC) codes that enable a limited number
of errors to be detected and corrected without retransmission. Block codes can be
used to improve the performance of a communications system when other means of
improvement (such as increasing transmitter power or using a more sophisticated
demodulator) are impractical.
In block codes, parity bits are added to blocks of message bits to make codewords
or code blocks. In a block encoder, k information bits are encoded into n code bits.
A total of n􀀀k redundant bits are added to the k information bits for the purpose of
detecting and correcting errors. The block code is referred to as an (n, k) code, and
the rate of the code is de ned as Rc = k=n and is equal to the rate of information
divided by the raw channel rate.
Parameters in Block Code
(a) Code Rate (Rc): As de ned above, Rc = k=n.
(b) Code Distance (d): Distance between two codewords is the number of ele-
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ments in which two codewords Ci and Cj di ers denoted by d (Ci;Cj ). If the code
used is binary, the distance is known as 'Hamming distance'. For example d(10110,
11011) is 3. If the code 'C' consists of the set of codewords, then the minimum
distance of the code is given by dmin = min fd (Ci;Cj)g.
(c) Code Weight (w): Weight of a codeword is given by the number of nonzero
elements in the codeword. For a binary code, the weight is basically the number of
1s in the codeword. For example weight of a code 101101 is 4.
Ex 1: The block code C = 00000, 10100, 11110, 11001 can be used to represent two
bit binary numbers as:
00 { 00000
01 { 10100
10 { 11110
11 { 11001
Here number of codewords is 4, k = 2, and n = 5.
To encode a bit stream 1001010011
First step is to break the sequence in groups of two bits, i.e., 10 01 01 00 11
Next step is to replace each block by its corresponding codeword, i.e.,
11110 10100 10100 00000 11001
Quite clearly, here, dmin = min fd (Ci;Cj)g = 2.
Properties of Block Codes
(a) Linearity: Suppose Ci and Cj are two code words in an (n, k) block code. Let
1 and 2 be any two elements selected from the alphabet. Then the code is said to
be linear if and only if 1C1 + 2C2 is also a code word. A linear code must contain
the all-zero code word.
(b) Systematic: A systematic code is one in which the parity bits are appended
to the end of the information bits. For an (n, k) code, the rst k bits are identical
to the information bits, and the remaining n 􀀀 k bits of each code word are linear
combinations of the k information bits.
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(c) Cyclic: Cyclic codes are a subset of the class of linear codes which satisfy the
following cyclic shift property: If C = [Cn􀀀1;Cn􀀀2; :::;C0] is a code word of a cyclic
code, then [Cn􀀀2;Cn􀀀3; :::;C0;Cn􀀀1], obtained by a cyclic shift of the elements of C,
is also a code word. That is, all cyclic shifts of C are code words.
In this context, it is important to know about Finite Field or Galois Field.
Let F be a nite set of elements on which two binary operations { addition (+) and
multiplication (:) are de ned. The set F together with the two binary operations is
called a eld if the following conditions are satis ed:
1. F is a commutative group under addition.
2. The set of nonzero elements in F is a commutative group under multiplication.
3. Multiplication is distributive over addition; that is, for any three elements a, b,
and c in F, a(b + c) = ab + ac
4. Identity elements 0 and 1 must exist in F satisfying a + 0 = a and a:1 = a.
5. For any a in F, there exists an additive inverse (􀀀a) such that a + (􀀀a) = 0.
6. For any a in F, there exists an multiplicative inverse a􀀀1 such that a:a􀀀1 = 1.
Depending upon the number of elements in it, a eld is called either a nite or an
in nite eld. The examples of in nite eld include Q (set of all rational numbers),
R (set of all real numbers), C (set of all complex numbers) etc. A eld with a nite
number of elements (say q) is called a 'Galois Field' and is denoted by GF(q). A
nite eld entity p(x), called a polynomial, is introduced to map all symbols (with
several bits) to the element of the nite eld. A polynomial is a mathematical
expression
p (x) = p0 + p1x + ::: + pmxm (7.31)
where the symbol x is called the indeterminate and the coe cients p0; p1; :::; pm are
the elements of GF(q). The coe cient pm is called the leading coe cient. If pm
is not equal to zero, then m is called the degree of the polynomial, denoted as deg
p(x). A polynomial is called monic if its leading coe cient is unity. The division
algorithm states that for every pair of polynomials a(x) and b(x) in F(x), there
exists a unique pair of polynomials q(x), the quotient, and r(x), the remainder,
such that a(x) = q(x)b(x) + r(x), where deg r(x)<deg b(x). A polynomial p(x) in
F(x) is said to be reducible if p(x)=a(x)b(x), otherwise it is called irreducible. A
monic irreducible polynomial of degree at least one is called a prime polynomial.
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An irreducible polynomial p(x) of degree `m' is said to be primitive if the smallest
integer `n' for which p(x) divides xn+1 is n = 2m􀀀1. A typical primitive polynomial
is given by p(x) = xm + x + 1.
A speci c type of code which obeys both the cyclic property as well as polynomial
operation is cyclic codes. Cyclic codes are a subset of the class of linear
codes which satisfy the cyclic property. These codes possess a considerable amount
of structure which can be exploited. A cyclic code can be generated by using a
generator polynomial g(p) of degree (n-k). The generator polynomial of an (n,k)
cyclic code is a factor of pn + 1 and has the form
g (p) = pn􀀀k + gn􀀀k􀀀1pn􀀀k􀀀1 + + g1p + 1: (7.32)
A message polynomial x(p) can also be de ned as
x (p) = xk􀀀1pk􀀀1 + + x1p + x0 (7.33)
where (xk􀀀1; : : : ; x0) represents the k information bits. The resultant codeword c(p)
can be written as
c (p) = x (p) g (p) (7.34)
where c(p) is a polynomial of degree less than n. We would see an application of
such codes in Reed-Solomon codes.
Examples of Block Codes
(a) Single Parity Check Code: In single parity check codes (example: ASCII code),
an overall single parity check bit is appended to 'k' information bits. Let the information
bit word be: (b1; b2; :::; bk), then parity check bit: p = b1 + b2 + ::::::::: + bk
modulo 2 is appended at the (k+1)th position, making the overall codeword: C =
(b1; b2; :::; bk; p). The parity bit may follow an even parity or an odd parity pattern.
All error patterns that change an odd number of bits are detectable, and all even
numbered error patterns are not detectable. However, such codes can only detect
the error, it cannot correct the error.
Ex. 2: Consider a (8,7) ASCII code with information codeword (0, 1, 0, 1, 1, 0, 0)
and encoded with overall even parity pattern. Thus the overall codeword is (0, 1, 0,
1, 1, 0, 0, 1) where the last bit is the parity bit. If there is a single error in bit 3: (0,
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1, 1, 1, 1, 0, 0, 1), then it can be easily checked by the receiver that now there are
odd number of 1's in the codeword and hence there is an error. On the other hand,
if there are two errors, say, errors in bit 3 and 5: (0, 1, 1, 1, 0, 0, 0, 1), then error
will not be detected.
After decoding a received codeword, let pc be the probability that the decoder
gives correct codeword C, pe is the probability that the decoder gives incorrect
codeword C0 6= C, and pf is the probability that the decoder fails to give a codeword.
In this case, we can write pc + pe + pf = 1.
If in an n-bit codeword, there are j errors and p is the bit error probability,
then the probability of obtaining j errors in this codeword is Pj = nCjpj (1 􀀀 p)n􀀀j .
Using this formula, for any (n; n 􀀀 1) single parity check block code, we get
pc = P0,
pe = P2 + P4 + ::: + P0
n (n0 = n if n is even, otherwise n0 = n 􀀀 1),
pf = P1 + P3 + ::: + P0
n (n0 = n 􀀀 1 if n is even, otherwise n0 = n).
As an example, for a (5,4) single parity check block code, pc = P0, pe = P2 + P4,
and pf = P1 + P3 + P5.
(b) Product Codes: Product codes are a class of linear block codes which provide
error detection capability using product of two block codes. Consider that nine
information bits (1; 0; 1; 0; 0; 1; 1; 1; 0) are to be transmitted. These 9 bits can be
divided into groups of three information bits and (4,3) single parity check codeword
can be formed with even parity. After forming three codewords, those can be appended
with a vertical parity bit which will form the fourth codeword. Thus the
following codewords are transmitted:
C1 = [1 0 1 0]
C2 = [0 0 1 1]
C3 = [1 1 0 0]
C4 = [0 1 0 1]:
Now if an error occurs in the second bit of the second codeword, the received codewords
at the receiver would then be
C1 = [1 0 1 0]
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C2 = [0 1 1 1]
C3 = [1 1 0 0]
C4 = [0 1 0 1]
"
and these would indicate the corresponding row and column position of the erroneous
bit with vertical and horizontal parity check. Thus the bit can be corrected. Here
we get a horizontal (4, 3) codeword and a vertical (4, 3) codeword and concatenating
them we get a (16, 9) product code. In general, a product code can be formed as
(n1; k1) & (n2; k2) ! (n1n2; k1k2).
(c) Repetition Codes: In a (n,1) repetition code each information bit is repeated
n times (n should be odd) and transmitted. At the receiver, the majority decoding
principle is used to obtain the information bit. Accordingly, if in a group of n received
bit, 1 occurs a higher number of times than 0, the information bit is decoded as 1.
Such majority scheme works properly only if the noise a ects less than n/2 number
of bits.
Ex 3: Consider a (3,1) binary repetition code.
For input bit 0, the codeword is (0 0 0) and for input bit 1, the codeword is
(1 1 1).
If the received codeword is (0 0 0), i.e. no error, it is decoded as 0.
Similarly, if the received codeword is (1 1 1), i.e. no error, it is decoded as 1.
If the received codeword is (0 0 1) or (0 1 0) or (1 0 0), then error is detected
and it is decoded as 0 with majority decoding principle.
If the received codeword is (0 1 1) or (1 1 0) or (1 0 1), once again error is
detected and it is decoded as 1 with majority decoding principle.
For such a (3,1) repetition code, pc = P0 + P1, pe = P2 + P3, and pf = 0.
(d) Hamming Codes: A binary Hamming code has the property that
(n; k) = (2m 􀀀 1; 2m 􀀀 1 􀀀 m) (7.35)
where k is the number of information bits used to form a n bit codeword, and m
is any positive integer. The number of parity symbols are n 􀀀 k = m. Thus, a
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codeword is represented by C = [i1; :::in; p1; :::; pn􀀀k]. This is quite a useful code in
communication which is illustrated via the following example.
Ex 4: Consider a (7, 4) Hamming code. With three parity bits we can correct exactly
1 error. The parity bits may follow such a modulo 2 arithmetic:
p1 = i1 + i2 + i3;
p2 = i2 + i3 + i4;
p3 = i1 + i3 + i4;
which is same as,
p1 + i1 + i2 + i3 = 0
p2 + i2 + i3 + i4 = 0
p3 + i1 + i3 + i4 = 0:
The transmitted codeword is then C = [i1; i2; :::; i4; p1; p2; p3].
Syndrome Decoding: For this Hamming code, let the received codeword be V =
[v1; v2; :::; v4; v5; v6; v7]. We de ne a syndrome vector S as
S = [S1 S2 S3]
S1 = v1 + v2 + v3 + v5
S2 = v2 + v3 + v4 + v6
S3 = v1 + v2 + v4 + v7
It is obvious that in case of no error, the syndrome vector is equal to zero. Corresponding
to this syndrome vector, there is an error vector e which can be obtained
from a syndrome table and nally the required codeword is taken as C = V + e. In
a nutshell, to obtain the required codeword, we perform the following steps:
1. Calculate S from decoder input V.
2. From syndrome table, obtain e corresponding to S.
3. The required codeword is then C = V + e.
A few cases are given below to illustrate the syndrome decoding.
1. Let C = [0 1 1 1 0 1 0] and V = [0 1 1 1 0 1 0]. This implies S = [0 0 0], and it
corresponds to e = [0 0 0 0 0 0 0]. Thus, C = V + e = [0 1 1 1 0 1 0].
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2. Let C = [1 1 0 0 0 1 0] and V = [1 1 0 1 0 1 0]. This means S = [0 1 1], from which
we get e = [0 0 0 1 0 0 0] which means a single bit error is there in the received bit
v4. This will be corrected by performing the operation C = V + e.
3. Another interesting case is, let C = [0 1 0 1 1 0 0] and V = [0 0 1 1 1 0 1] (two
errors at second and third bits). This makes S = [0 0 0] and as a result, e = [0 0
0 0 0 0 0]. However, C 6= V , and C = V + e implies the double error cannot be
corrected. Therefore a (7,4) Hamming code can correct only single bit error.
(e) Golay Codes: Golay codes are linear binary (23,12) codes with a minimum
distance of seven and a error correction capability of three bits. This is a special,
one of a kind code in that this is the only nontrivial example of a perfect code.
Every codeword lies within distance three of any codeword, thus making maximum
likelihood decoding possible.
(f) BCH Codes: BCH code is one of the most powerful known class of linear
cyclic block codes, known for their multiple error correcting ability, and the ease
of encoding and decoding. It's block length is n = 2m 􀀀 1 for m 3 and number
of errors that they can correct is bounded by t < (2m 􀀀 1)/2. Binary BCH codes
can be generalized to create classes of non binary codes which use m bits per code
symbol.
(g) Reed Solomon (RS) Codes: Reed-Solomon code is an important subset of
the BCH codes with a wide range of applications in digital communication and data
storage. Typical application areas are storage devices (CD, DVD etc.), wireless
communications, digital TV, high speed modems. It's coding system is based on
groups of bits, such as bytes, rather than individual 0 and 1. This feature makes it
particularly good at dealing with burst of errors: six consecutive bit errors. Block
length of these codes is n = 2m 􀀀 1, and can be extended to 2m or 2m + 1. Number
of parity symbols that must be used to correct e errors is n 􀀀 k = 2e. Minimum
distance dmin = 2e + 1, and it achieves the largest possible dmin of any linear code.
For US-CDPD, the RS code is used with m = 6. So each of the 64 eld elements
is represented by a 6 bit symbol. For this case, we get the primitive polynomial as
p(x) = x6 + x + 1. Equating p(x) to 0 implies x6 = x + 1.
The 6 bit representation of the nite eld elements is given in Table 7.1. The table
elements continue up to 62. However, to follow linearity property there should be
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Table 7.1: Finite eld elements for US-CDPD
5 4 3 2 1 0
1 0 0 0 0 0 1
1 0 0 0 1 0 0
2 0 0 1 0 0 0
. . . . . . .
. . . . . . .
6 = + 1 0 0 0 0 1 1
. . . . . . .
. . . . . . .
a zero codeword, hence 63 is assigned zero.
The encoding part of the RS polynomial is done as follows:
Information polynomial: d(x) = Cn􀀀1xn􀀀1 + Cn􀀀2xn􀀀2 + ::::: + C2tx2t,
Parity polynomial: p(x) = C2t􀀀1x2t􀀀1 + ::: + C0,
Codeword polynomial: c(x) = d(x) + p(x).
Since generating an information polynomial is di cult, so a generating polynomial is
used instead. Information polynomial is then the multiple of generating polynomial.
This process is given below.
Since this kind of codes are cyclic codes, we take a generating polynomial g(x)
such that d(x) = g(x)q(x) + r(x) where q(x) is the quotient polynomial and r(x)
is the remainder polynomial. The codeword polynomial would then be given as:
c(x) = g(x)q(x) + r(x) = p(x). If we assign a parity polynomial p(x) = r(x), then
the codeword polynomial c(x) = g(x)p(x) and the entire process becomes easier.
On the decoder side one has to nd a speci c r(x) = p(x) or vice-versa, but due
to its complexity, it is mainly done using syndrome calculation. The details of such
a syndrome calculation can be found in [1].
7.4.3 Convolutional Codes
A continuous sequence of information bits is mapped into a continuous sequence of
encoder output bits. A convolutional code is generated by passing the information
sequence through a nite state shift register. Shift register contains 'N' k-bit stages
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Figure 7.6: A convolutional encoder with n=2 and k=1.
and m linear algebraic function generators based on the generator polynomials.
Input data is shifted into and along the shift register, k-bits at a time. Number
of output bits for each k-bit user input data sequence is n bits, so the code rate
Rc = k=n. The shift register of the encoder is initialized to all-zero-state before
Figure 7.7: State diagram representation of a convolutional encoder.
encoding operation starts. It is easy to verify that encoded sequence is 00 11 10
00 01 : : : for an input message sequence of 01011 : : :. Convolution codes may be
represented in various ways as given below.
State Diagram:
Since the output of the encoder is determined by the input and the current
state of the encoder, a state diagram can be used to represent the encoding process.
The state diagram is simply a graph of the possible states of the encoder and the
possible transitions from one state to another. The path information between the
states, denoted as b/c1c2, represents input information bit 'b' and the corresponding
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Figure 7.8: Tree diagram representation of a convolutional encoder.
output bits (c1c2). Again, it is not di cult to verify from the state diagram that an
input information sequence b = (1011) generates an encoded sequence c = (11, 10,
00, 01).
Tree Diagram:
The tree diagram shows the structure of the encoder in the form of a tree with the
branches representing the various states and the outputs of the coder. The encoded
bits are labeled on the branches of the tree. Given an input sequence, the encoded
sequence can be directly read from the tree. As an example, an input sequence
(1011) results in the encoded sequence (11, 10, 00, 01).
Figure 7.9: Trellis diagram of a convolutional encoder.
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Figure 7.10: Block diagram of a turbo encoder.
Trellis Diagram:
Tree reveals that the structure repeats itself once the number of stages is greater
than the constraint length. It is observed that all branches emanating from two
nodes having the same state are identical in the sense that they generate identical
output sequences. This means that the two nodes having the same label can be
merged. By doing this throughout the tree diagram, we obtain another diagram
called a Trellis Diagram which is more compact representation.
7.4.4 Concatenated Codes
Concatenated codes are basically concatenation of block and convolutional codes. It
can be of two types: serial and parallel codes. Below, we discuss a popular parallel
concatenated code, namely, turbo code.
Turbo Codes: A turbo encoder is built using two identical convolutional codes
of special type with parallel concatenation. An individual encoder is termed a component
encoder. An interleaver separates the two component encoders. The interleaver
is a device that permutes the data sequence in some predetermined manner.
Only one of the systematic outputs from the two component encoders is used to form
a codeword, as the systematic output from the other component encoder is only a
permuted version of the chosen systematic output. Figure 7.10 shows the block diagram
of a turbo encoder using two identical encoders. The rst encoder outputs
the systematic V0 and recursive convolutional V1 sequences while the second encoder
discards its systematic sequence and only outputs the recursive convolutional
V2 sequence. Depending on the number of input bits to a component encoder it
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may be binary or m-binary encoder. Encoders are also categorized as systematic
or non-systematic. If the component encoders are not identical then it is called an
asymmetric turbo code.
7.5 Conclusion
Although a lot of advanced powerful techniques for mitigating the fading e ects
such as space diversity in MIMO systems, space-time block coding scheme, MIMO
equalization, BLAST architectures etc. have taken place in modern wireless communication,
nevertheless, the discussed topics in this chapter are the basic building
blocks for all such techniques and that stems the necessity for all these discussions.
The e ectiveness of the discussed topics would be more clear in the next chapter in
the context of di erent multiple access techniques.
7.6 References
1. T. S. Rappaport, Wireless Communications: Principles and Practice, 2nd ed.
Singapore: Pearson Education, Inc., 2002.
2. J. R. Treichler, C. R. Johnson (Jr.) and M. G. Larimore, Theory and Design
of Adaptive Filters. New Delhi: PHI, 2002.
3. S. Gravano, Introduction to Error Control Codes. NY: Oxford University
Press, 2001.
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Chapter 8
Multiple Access Techniques
Multiple access techniques are used to allow a large number of mobile users to share
the allocated spectrum in the most e cient manner. As the spectrum is limited, so
the sharing is required to increase the capacity of cell or over a geographical area
by allowing the available bandwidth to be used at the same time by di erent users.
And this must be done in a way such that the quality of service doesn't degrade
within the existing users.
8.1 Multiple Access Techniques for Wireless Communi-
cation
In wireless communication systems it is often desirable to allow the subscriber to
send simultaneously information to the base station while receiving information from
the base station.
A cellular system divides any given area into cells where a mobile unit in each
cell communicates with a base station. The main aim in the cellular system design
is to be able to increase the capacity of the channel i.e. to handle as many calls
as possible in a given bandwidth with a su cient level of quality of service. There
are several di erent ways to allow access to the channel. These includes mainly the
following:
1) Frequency division multiple-access (FDMA)
2) Time division multiple-access (TDMA)
3) Code division multiple-access (CDMA)
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Table 8.1: MA techniques in di erent wireless communication systems
Advanced Mobile Phone Systems: FDMA/FDD
Global System for Mobile: TDMA/FDD
U.S. Digital Cellular: TDMA/FDD
Japanese Digital Cellular: TDMA/FDD
CT2 Cordless Telephone: FDMA/TDD
Digital European Cordless Telephone: FDMA/TDD
U.S. Narrowband Spread Spectrum (IS-95): CDMA/FDD
4) Space Division Multiple access (SDMA)
FDMA,TDMA and CDMA are the three major multiple access techniques that
are used to share the available bandwidth in a wireless communication system.
Depending on how the available bandwidth is allocated to the users these techniques
can be classi ed as narrowband and wideband systems.
8.1.1 Narrowband Systems
The term narrowband is used to relate the bandwidth of the single channel to the
expected coherence bandwidth of the channel. The available spectrum is divided in
to a large number of narrowband channels. The channels are operated using FDD.
In narrow band FDMA, a user is assigned a particular channel which is not shared by
other users in the vicinity and if FDD is used then the system is called FDMA/FDD.
Narrow band TDMA allows users to use the same channel but allocated a unique
time slot to each user on the channel, thus separating a small number of users in time
on a single channel. For narrow band TDMA, there generally are a large number of
channels allocated using either FDD or TDD, each channel is shared using TDMA.
Such systems are called TDMA/FDD and TDMA/TDD access systems.
8.1.2 Wideband Systems
In wideband systems, the transmission bandwidth of a single channel is much larger
than the coherence bandwidth of the channel. Thus, multipath fading doesnt greatly
a ect the received signal within a wideband channel, and frequency selective fades
occur only in a small fraction of the signal bandwidth
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Figure 8.1: The basic concept of FDMA.
8.2 Frequency Division Multiple Access
This was the initial multiple-access technique for cellular systems in which each
individual user is assigned a pair of frequencies while making or receiving a call as
shown in Figure 8.1. One frequency is used for downlink and one pair for uplink.
This is called frequency division duplexing (FDD). That allocated frequency pair
is not used in the same cell or adjacent cells during the call so as to reduce the co
channel interference. Even though the user may not be talking, the spectrum cannot
be reassigned as long as a call is in place. Di erent users can use the same frequency
in the same cell except that they must transmit at di erent times.
The features of FDMA are as follows: The FDMA channel carries only one
phone circuit at a time. If an FDMA channel is not in use, then it sits idle and
it cannot be used by other users to increase share capacity. After the assignment
of the voice channel the BS and the MS transmit simultaneously and continuously.
The bandwidths of FDMA systems are generally narrow i.e. FDMA is usually
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implemented in a narrow band system The symbol time is large compared to the
average delay spread. The complexity of the FDMA mobile systems is lower than
that of TDMA mobile systems. FDMA requires tight ltering to minimize the
adjacent channel interference.
8.2.1 FDMA/FDD in AMPS
The rst U.S. analog cellular system, AMPS (Advanced Mobile Phone System) is
based on FDMA/FDD. A single user occupies a single channel while the call is in
progress, and the single channel is actually two simplex channels which are frequency
duplexed with a 45 MHz split. When a call is completed or when a hando occurs
the channel is vacated so that another mobile subscriber may use it. Multiple or
simultaneous users are accommodated in AMPS by giving each user a unique signal.
Voice signals are sent on the forward channel from the base station to the mobile
unit, and on the reverse channel from the mobile unit to the base station. In AMPS,
analog narrowband frequency modulation (NBFM) is used to modulate the carrier.
8.2.2 FDMA/TDD in CT2
Using FDMA, CT2 system splits the available bandwidth into radio channels in the
assigned frequency domain. In the initial call setup, the handset scans the available
channels and locks on to an unoccupied channel for the duration of the call. Using
TDD(Time Division Duplexing ), the call is split into time blocks that alternate
between transmitting and receiving.
8.2.3 FDMA and Near-Far Problem
The near-far problem is one of detecting or ltering out a weaker signal amongst
stronger signals. The near-far problem is particularly di cult in CDMA systems
where transmitters share transmission frequencies and transmission time. In contrast,
FDMA and TDMA systems are less vulnerable. FDMA systems o er di erent
kinds of solutions to near-far challenge. Here, the worst case to consider is recovery
of a weak signal in a frequency slot next to strong signal. Since both signals are
present simultaneously as a composite at the input of a gain stage, the gain is set
according to the level of the stronger signal; the weak signal could be lost in the
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noise
oor. Even if subsequent stages have a low enough noise
oor to provide
8.3 Time Division Multiple Access
In digital systems, continuous transmission is not required because users do not
use the allotted bandwidth all the time. In such cases, TDMA is a complimentary
access technique to FDMA. Global Systems for Mobile communications (GSM) uses
the TDMA technique. In TDMA, the entire bandwidth is available to the user but
only for a nite period of time. In most cases the available bandwidth is divided
into fewer channels compared to FDMA and the users are allotted time slots during
which they have the entire channel bandwidth at their disposal, as shown in Figure
8.2.
TDMA requires careful time synchronization since users share the bandwidth in
the frequency domain. The number of channels are less, inter channel interference
is almost negligible. TDMA uses di erent time slots for transmission and reception.
This type of duplexing is referred to as Time division duplexing(TDD).
The features of TDMA includes the following: TDMA shares a single carrier frequency
with several users where each users makes use of non overlapping time slots.
The number of time slots per frame depends on several factors such as modulation
technique, available bandwidth etc. Data transmission in TDMA is not continuous
but occurs in bursts. This results in low battery consumption since the subscriber
transmitter can be turned OFF when not in use. Because of a discontinuous transmission
in TDMA the hando process is much simpler for a subscriber unit, since it
is able to listen to other base stations during idle time slots. TDMA uses di erent
time slots for transmission and reception thus duplexers are not required. TDMA
has an advantage that is possible to allocate di erent numbers of time slots per
frame to di erent users. Thus bandwidth can be supplied on demand to di erent
users by concatenating or reassigning time slot based on priority.
8.3.1 TDMA/FDD in GSM
As discussed earlier, GSM is widely used in Europe and other parts of the world.
GSM uses a variation of TDMA along with FDD. GSM digitizes and compresses
data, then sends it down a channel with two other streams of user data, each in its
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Figure 8.2: The basic concept of TDMA.
own time slot. It operates at either the 900 MHz or 1800 MHz frequency band. Since
many GSM network operators have roaming agreements with foreign operators, users
can often continue to use their mobile phones when they travel to other countries.
8.3.2 TDMA/TDD in DECT
DECT is a pan European standard for the digitally enhanced cordless telephony
using TDMA/TDD. DECT provides 10 FDM channels in the band 1880-1990 Mhz.
Each channel supports 12 users through TDMA for a total system load of 120 users.
DECT supports handover, users can roam over from cell to cell as long as they remain
within the range of the system. DECT antenna can be equipped with optional spatial
diversity to deal with multipath fading.
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8.4 Spread Spectrum Multiple Access
Spread spectrum multiple access (SSMA) uses signals which have a transmission
bandwidth whose magnitude is greater than the minimum required RF bandwidth.
A pseudo noise (PN) sequence converts a narrowband signal to a wideband noise
like signal before transmission. SSMA is not very bandwidth e cient when used
by a single user. However since many users can share the same spread spectrum
bandwidth without interfering with one another, spread spectrum systems become
bandwidth e cient in a multiple user environment.
There are two main types of spread spectrum multiple access techniques: Frequency
hopped multiple access (FHMA)
Direct sequence multiple access (DSMA) or
Code division multiple access (CDMA).
8.4.1 Frequency Hopped Multiple Access (FHMA)
This is a digital multiple access system in which the carrier frequencies of the individual
users are varied in a pseudo random fashion within a wideband channel.
The digital data is broken into uniform sized bursts which is then transmitted on
di erent carrier frequencies.
8.4.2 Code Division Multiple Access
In CDMA, the same bandwidth is occupied by all the users, however they are all
assigned separate codes, which di erentiates them from each other (shown in Figure
8.3). CDMA utilize a spread spectrum technique in which a spreading signal (which
is uncorrelated to the signal and has a large bandwidth) is used to spread the narrow
band message signal.
Direct Sequence Spread Spectrum (DS-SS)
This is the most commonly used technology for CDMA. In DS-SS, the message signal
is multiplied by a Pseudo Random Noise Code. Each user is given his own codeword
which is orthogonal to the codes of other users and in order to detect the user, the
receiver must know the codeword used by the transmitter. There are, however, two
problems in such systems which are discussed in the sequel.
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Figure 8.3: The basic concept of CDMA.
CDMA/FDD in IS-95
In this standard, the frequency range is: 869-894 MHz (for Rx) and 824-849 MHz
(for Tx). In such a system, there are a total of 20 channels and 798 users per channel.
For each channel, the bit rate is 1.2288 Mbps. For orthogonality, it usually combines
64 Walsh-Hadamard codes and a m-sequence.
8.4.3 CDMA and Self-interference Problem
In CDMA, self-interference arises from the presence of delayed replicas of signal due
to multipath. The delays cause the spreading sequences of the di erent users to
lose their orthogonality, as by design they are orthogonal only at zero phase o set.
Hence in despreading a given user's waveform, nonzero contributions to that user's
signal arise from the transmissions of the other users in the network. This is distinct
from both TDMA and FDMA, wherein for reasonable time or frequency guardbands,
respectively, orthogonality of the received signals can be preserved.
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8.4.4 CDMA and Near-Far Problem
The near-far problem is a serious one in CDMA. This problem arises from the fact
that signals closer to the receiver of interest are received with smaller attenuation
than are signals located further away. Therefore the strong signal from the nearby
transmitter will mask the weak signal from the remote transmitter. In TDMA and
FDMA, this is not a problem since mutual interference can be ltered. In CDMA,
however, the near-far e ect combined with imperfect orthogonality between codes
(e.g. due to di erent time sifts), leads to substantial interference. Accurate and fast
power control appears essential to ensure reliable operation of multiuser DS-CDMA
systems.
8.4.5 Hybrid Spread Spectrum Techniques
The hybrid combinations of FHMA, CDMA and SSMA result in hybrid spread
spectrum techniques that provide certain advantages. These hybrid techniques are
explained below,
Hybrid FDMA/CDMA (FCDMA):
An alternative to the CDMA technique in which the available wideband spectrum
is divided into a smaller number of sub spectra with smaller bandwidths. The smaller
sub channels become narrow band CDMA systems with processing gain lower than
the original CDMA system. In this scheme the required bandwidth need not be
contiguous and di erent user can be allotted di erent sub spectrum bandwidths
depending on their requirements. The capacity of this hybrid FCDMA technique is
given by the sum of the capacities of a system operating in the sub spectra.
Hybrid Direct Sequence/Frequency Hopped Multiple Access Techniques (DS/FHMA):
A direct sequence modulated signal whose center frequency is made to hop periodically
in a pseudo random fashion is used in this technique. One of the advantages
using this technique is they avoid near-far e ect. However, frequency hopped
CDMA systems are not adaptable to the soft hando process since it is di cult
to synchronize the frequency hopped base station receiver to the multiple hopped
signals. Time and Code Division Multiple Access (TCDMA):
In this TCDMA method di erent cells are allocated di erent spreading codes.
In each cell, only one user per cell is allotted a particular time slot. Thus at any
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time only one user is transmitting in each cell. When a hando takes place the
spreading code of that user is changed to the code of the new cell. TCDMA also
avoids near-far e ect as the number of users transmitting per cell is one.
Time Division Frequency Hopping (TDFH):
This technique has been adopted for the GSM standard, where the hopping sequence
is prede ned and the subscriber is allowed to hop only on certain frequencies
which are assigned to a cell. The subscriber can hop to a new frequency at the start
of a new TDMA frame, thus avoiding a severe fade or erasure event on a particular
channel. This technique has the advantage in severe multipath or when severe
channel interference occurs.
8.5 Space Division Multiple Access
SDMA utilizes the spatial separation of the users in order to optimize the use of the
frequency spectrum. A primitive form of SDMA is when the same frequency is reused
in di erent cells in a cellular wireless network. The radiated power of each user
is controlled by Space division multiple access. SDMA serves di erent users by using
spot beam antenna. These areas may be served by the same frequency or di erent
frequencies. However for limited co-channel interference it is required that the cells
be su ciently separated. This limits the number of cells a region can be divided into
and hence limits the frequency re-use factor. A more advanced approach can further
increase the capacity of the network. This technique would enable frequency re-use
within the cell. In a practical cellular environment it is improbable to have just one
transmitter fall within the receiver beam width. Therefore it becomes imperative
to use other multiple access techniques in conjunction with SDMA. When di erent
areas are covered by the antenna beam, frequency can be re-used, in which case
TDMA or CDMA is employed, for di erent frequencies FDMA can be used.
8.6 Conclusion
In this chapter, we have mainly discussed the xed assignment type of MA techniques,
namely, FDMA, TDMA and CDMA. We have, however, intensionally not
covered the reservation-based MA schemes such as packet reservation MA or polling
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or token passing etc. The main idea to discuss only the basic MA techniques has
been to grow up a fair idea about the resource sharing in a wireless media when there
are many users, keeping the QoS view point in mind. The readers are encouraged
to go through the advanced topics once they nish reading the discussed 8 chapters
in this lecture notes.
8.7 References
1. T. S. Rappaport, Wireless Communications: Principles and Practice, 2nd ed.
Singapore: Pearson Education, Inc., 2002.
2. K. Feher, Wireless Digital Communications: Modulation and Spread Spectrum
Applications. Upper Saddle River, NJ: Prentice Hall, 1995.
3. J. G. Proakis, Digital Communications, 4th ed. NY: McGraw Hill, 2000.
4. G. R. Cooper and C. D. McGillem, Modern Communications and Spread Spec-
trum, NY: McGraw Hill, 1986.
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